Fairchild SEMICONDUCTOR RC5040 User Manual

www.fairchildsemi.com

Application Note 42

Implementing the RC5040 and RC5042

DC-DC Converters on Pentium^®Pro Motherboards

Introduction

PentiumPro and OverDrive^®

Processor Power Requirements

This document describes how to implement a switching volt-

age regulator using an RC5040 or an RC5042 high speed

controller, a power inductor, a Schottky diode, appropriate

capacitors, and external power MOSFETs. This regulator

forms a step down DC-DC converter that can deliver up to

14.5A of continuous load current at voltages ranging from

2.1V to 3.5V. A speciﬁc application circuit, design consider-

ations, component selection, PCB layout guidelines and per-

formance evaluation procedures are covered in detail.

Use Intel’s AP-523 Application Note, Pentium® Pro

Processor Power Distribution Guidelines, November 1995

(order number 242764-001), as a basic reference. The speci-

ﬁcations contained in this document have been modiﬁed

slightly from the original Intel document to include updated

speciﬁcations for Pentium Pro microprocessors. Please con-

tact Intel Corporation for speciﬁc details.

Input Voltages

In the past 10 years, microprocessors have evolved at such an

exponential rate that a modern chip can rival the computing

power of a mainframe computer. Such evolution has been

possible because of the increasing numbers of transistors that

processors integrate. Pentium CPUs, for example, integrate

well over 5 million transistors on a single piece of silicon.

Available inputs are +5V ±5% and +12V ±5%. Raytheon

Electronics’ DC-DC converters may use either or both

inputs. Their input voltage requirements are listed in Table 1.

Table 1. Input Voltage Requirements

Controller

MOSFET

Drain

MOSFET

Gate Bias

To integrate so many transistors on a piece of silicon, their

physical geometry has been reduced to the sub-micron level.

As a result of each geometry reduction, the corresponding

operational voltage for each transistor has also been reduced.

This changing voltage for the CPU demands the design of a

programmable power supply—a design that is not com-

pletely re-engineered with every change in CPU voltage.

Part #

RC5040

RC5042

+5V ±5%

+5V ±5% or

12V ±5%

RC5043

+5V ±5%

12V ±5%

Pentium Pro DC Power Requirements

Refer to Table 2 for the power supply speciﬁcations for

Pentium Pro and Overdrive Processors. For a motherboard

design without a standard Voltage Regulator Module (VRM)

socket, the on-board DC-DC converter must supply a mini-

mum I P current of 13.9A at 2.5V and 12.4A at 3.3V. For a

ﬂexible motherboard design, the on-board converter must be

The operational voltage of CPUs has shown a downwards

trend for the past 5 years: from 5V for the x386 and x486, to

3.3V for Pentium, and 3.1V for Pentium Pro. Furthermore,

emerging chip technologies may require operating voltages

as low as 2.5V. With this trend in mind, Raytheon Electron-

ics has designed the RC5040 and RC5042 controllers. These

controllers integrate the necessary programmability to

address the changing power supply requirements of lower

voltage CPUs.

able to supply 14.5A maximum I P.

DC Voltage Regulation

As indicated in Table 2, the voltage level supplied to the

CPU must be within ±5% of its nominal setting. Voltage

regulation limits must include:

Previous generations of DC-DC converter controllers were

designed with ﬁxed output voltages adjustable only with a

set of external resistors. In a high volume production envi-

ronment (such as with personal computers), however, a CPU

voltage change requires a CPU board re-design to accommo-

date the new voltage requirement. The integrated 4-bit DAC

in the RC5040 and the RC5042 reads the voltage ID code

from the Pentium Pro microprocessor and conﬁgures the sys-

tem to provide the appropriate voltage. In this manner, the

PC board does not have to be re-designed each time the CPU

voltage changes. The CPU can thus automatically conﬁgure

its own required voltage.

• Output load ranges speciﬁed in Table 2

• Output ripple/noise

• DC output initial voltage set point

• Temperature and warm up drift (Ambient +10°C to +60°C

at full load with a maximum rate of change of 5°C per 10

minutes minimum but no more than 10°C per hour)

• Output load transient with:

Slew rate >30A/µs at the converter pins

Range: 0.3A – I P Max (as deﬁned in Table 2).

Rev. 1.1.0

APPLICATION NOTE

AN42

The RC5040 and RC5042 Controllers

RC5040 and RC5042 Description

The RC5040 is a programmable synchronous-mode DC-DC

converter controller. The RC5042 is a non-synchronous ver-

sion of the RC5040. When designed with the appropriate

external components, either device can be conﬁgured to

deliver more than 14.5A of output current. During heavy

loading conditions, these controllers function as current-

mode PWM step-down regulators. Under light loads, they

function in PFM (pulse frequency modulation) or pulse skip-

ping mode. The controllers sense the load level and switch

between the two operating modes automatically, thus opti-

mizing efﬁciency under all loads. The key differences

between the RC5040 and RC5042 are listed in Table 4.

Simple Step-Down Converter

V_IN

R_LVout

–

65-AP42-01

Figure 1. Simple Buck DC-DC Converter

Figure 1 illustrates a step-down DC-DC converter with no

feedback control. The basic step-down converter serves as

the basis for deriving the design equations for the RC5040

and RC5042. From Figure 1, the basic operation begins by

closing the switch S1, so that the input voltage V is

impressed across inductor L1. The current ﬂowing through

this inductor is given by the following equation:

Table 4. RC5040 and RC5042 Differences

RC5040

RC5042

Operation

Package

Synchronous Non-Synchronous

20-pin SOIC

Yes

16-pin SOIC

Output Enable/

Disable

(V – V

OUT ON

= -----------------------------------------------

Refer to the RC5040 Block Diagram illustrated in Figure 2.

The control loop of the regulator contains two main sections:

the analog control block and the digital control block. The

analog block consists of signal conditioning ampliﬁers feed-

ing into a set of comparators which provide the inputs to the

digital block. The signal conditioning section accepts inputs

from the IFB (current feedback) and VFB (voltage feedback)

pins and sets two controlling signal paths. The voltage con-

trol path ampliﬁes the VFB signal and presents the output to

one of the summing ampliﬁer inputs. The current control

path takes the difference between the IFB and VFB and pre-

sents the result to another input of the summing ampliﬁer.

These two signals are then summed together with the slope

compensation input from the oscillator. This output is then

presented to a comparator, which provides the main PWM

control signal to the digital control block.

where T is the duty cycle (the time when S1 is closed).

When S1 opens, the diode D1 conducts the inductor

current and the output current is delivered to the load accord-

ing to the following equation:

(T – T

)

OUT

= -------------------------------------------

where T is the overall switching period and (T – T ) is

the time during which S1 is open.

By solving these equations you can obtain the basic relation-

ship for the output voltage of a step-down converter:

----------

= V

OUT

The additional comparators in the analog control section sets

the threshold for when the RC5040 enters PFM mode during

light loads and the point when the current limit comparator

disables the output drive signals to the MOSFETs.

In order to obtain a more accurate approximation for V

OUT

we must also include the forward voltage V across diode

D1 and the switching loss, V . After taking into account

these factors, the new relationship becomes:

The digital control block is designed to take the comparator

inputs along with the main clock signal from the oscillator

and provide the appropriate pulses to the HIDRV and

LODRV pins that control the external power MOSFETs. The

digital section was designed utilizing high speed Schottky

transistor logic, thus allowing the RC5040 to operate at clock

speeds as high as 1MHz.

----------

= (V + V – V

)

– V

OUT

Where V

= I • R .

DS,ON

AN42

APPLICATION NOTE

Main Control Loop

RC5040

+5V

VIN

OSCILLATOR

–

DIGITAL

CONTROL

1.24V

REFERENCE

4-BIT

DAC

VREF

POWER

GOOD

PWRGD

65-5040-01

VID0

VID1

VID2

VID3

Figure 2. RC5040 Block Diagram

High Current Output Drivers

Power Good (PWRGD)

The RC5040 contains two identical high current output

drivers that use high speed bipolar transistors in a push-pull

conﬁguration. Each driver is capable of delivering 1A of cur-

rent in less than 100ns. Each driver’s power and ground are

separated from the chip power and ground for additional

switching noise immunity. The HIDRV driver’s power sup-

ply, VCCQP, is boot-strapped from a ﬂying capacitor as

illustrated in Figure 3. Using this conﬁguration, C12 is

charged from VCC via the Schottky diode DS2 and boosted

when the FET is turned on. This scheme provides a VCCQP

voltage equal to 2•VCC – VDS(DS2), or approximately 9.5V

when VCC = 5V. This voltage is sufﬁcient to provide the 9V

gate drive to the MOSFET that is required to achieve a low

R^DS(ON). Since the low side synchronous FET is referenced to

ground (see Figure 4), boosting the gate drive voltage is not

needed and the VCCP power pin can be tied to VCC.

Refer to Typical Operating Characteristics of the RC5040

data sheet for a full load VCCQP waveform.

The RC5040 and RC5042 Power Good function has been

designed according to Intel’s Pentium Pro DC-DC converter

speciﬁcation. The Power Good function provides a constant

voltage monitor on the VFB pin. The internal circuitry of the

converter compares the VFB signal to the VREF voltage and

outputs an active-low interrupt signal to the CPU when the

power supply voltage exceeds ±7% of its nominal setpoint.

The Power Good ﬂag provides no other control function to

the RC5040.

Output Enable (OUTEN)

Intel speciﬁcations state that the DC-DC converter should

accept an open collector signal for controlling the output

voltage. A logic LOW for this signal disables the output volt-

age. When disabled, the PWRGD output is in the low state.

This feature is available for the RC5040 only.

Upgrade Present (UP#)

Intel speciﬁcations state that the DC-DC converter must

accept an open collector signal that indicates the presence of

an upgrade processor. The typical state is high (for a stan-

dard P6 processor). When the signal is low or in theground

state (for the OverDrive processor), the output voltage must

be disabled unless the converter can supply the OverDrive

processor’s power requirements. When disabled, the

PWRGD output must be in the low state. Because the

RC5040 and RC5042 can supply the OverDrive processor

requirements, the UP# signal is not required.

Internal Voltage Reference

The reference used in the RC5040 is a precision band-gap

voltage reference, with internal resistors precisely trimmed

to provide a near zero temperature coefﬁcient, TC. Added to

the reference voltage is the output from a 4-bit DAC. The

DAC is provided meet Pentium Pro speciﬁcations, requiring

a programmable converter output via a 4-bit voltage identiﬁ-

cation (VID) code. This code scales the output voltage from

2.0V (no CPU) to 3.5V in 100mV increments. To guarantee

stable operation under all loads, a 10KΩ pull-up resistor and

0.1µF of decoupling capacitance should be connected to the

VREF pin. No load should be imposed on this pin.

APPLICATION NOTE

AN42

In general, a lower operating frequency increases the peak

Over-Voltage Protection

ripple current ﬂowing through the output inductor, allowing

the use of a larger inductor value. Operation at lower fre-

quencies increases the amount of energy storage that the

bulk output capacitors must provide during load transients

that occur due to the slower loop response of the controller.

The RC5040 and RC5042 constantly monitor the output

voltage for protection against over voltage. If the voltage at

the VFB pin exceeds 20% of the selected program voltage,

an over-voltage condition is assumed, and the controller dis-

ables the output drive signal to the external MOSFET(s).

In addition, note that the efﬁciency losses due to switching

are relatively ﬁxed per switching cycle. Therefore, as the

switching frequency increases, the contribution toward efﬁ-

ciency due to switching losses also increases.

Short Circuit Protection

A current sense methodology is implemented to disable the

output drive signal to the MOSFET(s) when an over-current

condition is detected. The voltage drop created by the output

current ﬂowing across a sense resistor is presented to an

internal comparator. When the voltage developed across the

sense resistor exceeds the comparator threshold voltage,

the controller disables the output drive signal to the

MOSFET(s).

RC5040 has an optimal operating frequency of 650KHz.

This frequency allows the use of smaller inductive and

capacitive components while optimizing peak efﬁciency

under all operating conditions.

Design Considerations and

Component Selection

The DC-DC converter returns to normal operation after the

fault has been removed, for either an over voltage or a short

circuit condition.

Application Circuits

Oscillator

Figure 3 illustrates a typical non-synchronous application

using the RC5040. Figure 4 shows a typical synchronous

application using the RC5040, and Figure 5 shows a typical

non-synchronous application using the RC5042.

The RC5040 oscillator section is implemented using a

ﬁxed current capacitor charging conﬁguration. An external

capacitor (CEXT) is used to preset the oscillator frequency

between 200KHz and 1MHz. This allows maximum ﬂexibil-

ity in setting the switching frequency and in choosing exter-

nal components.

VCC

2.6µH

0.1µF

1000µF 1000µF

1000µF

DS2

1N5817

0.1µF

C12

2SK1388

1µF

2SK1388

SENSE

10K

1.3µH

8mΩ

RC5040

4.7µF

VREF

GND

DS1

MBR1545CT

0.1µF

EXT

39pF

10K

VID3

VID2

VCC

10K

PWRGD

VID1

VID0

65-AP42-03

C11

0.22µF

VCC

10K

OUTEN

C10

0.1µF

Figure 3. Non-Synchronous DC-DC Converter Application Schematic Using RC5040

AN42

APPLICATION NOTE

VCC

2.6µH

0.1µF

1000µF

DS2

1N5817

0.1µF

C12

2SK1388

1µF

2SK1388

SENSE

10K

1.3µH

8mΩ

4.7µF

RC5040

VREF

2SK1388

DS1

1N5817

0.1µF

GND

EXT

39pF

10K

VID3

VCC

VID2

10K

PWRGD

VID1

VID0

65-AP42-04

C11

VCC

0.22µF

10K

OUTEN

C10

0.1µF

Figure 4. Synchronous DC-DC Converter Application Schematic Using RC5040

2.6µH

VCC

0.1µF

1000µF 1000µF

1000µF

DS2

1N5817

0.1µF

C12

2SK1388

1µF

2SK1388

SENSE

1.3µH

10K

8mΩ

4.7µF

VREF

GND

RC5042

DS1

MBR1545CT

0.1µF

EXT

39pF

10K

VID3

VCC

10K

VID2

65-AP42-05

PWRGD

VID1

VID0

C11

0.22µF

VCC

C10

0.1µF

Figure 5. Non-Synchronous DC-DC Converter Application Schematic Using RC5042

APPLICATION NOTE

AN42

• Power package with low thermal resistance

• Drain current rating of 20A minimum

• Drain-Source voltage > 15V.

MOSFET Selection

This application requires the use of N-channel, Logic Level

Enhancement Mode Field Effect Transistors. The desired

characteristics of these components are:

The on-resistance (R ) is the main parameter for MOS-

DS,ON

FET selection. It determines the MOSFET’s power dissipa-

tion, thus signiﬁcantly affecting the efﬁciency of the

converter. Several suitable MOSFETs are shown in Table 5.

• Low Static Drain-Source On-Resistance

< 37 mΩ (lower is better)

DS,ON

• Low gate drive voltage, V ≤ 4.5V

Table 5. MOSFET Selection Table

DS,ON

(mΩ)

Thermal

Manufacturer & Model #

Conditions

T = 25°C

Typ.

Max.

P ackage Resistance

Fuji

2SK1388

= 4V

—

TO-220

= 75

= 50

I = 17.5A

T = 125°C

Siliconix

SI4410DY

= 4.5V

T = 25°C

16.5

SO-8

(SMD)

I = 5A

T = 125°C

National Semiconductor

NDP706AL

= 5V

T = 25°C

TO-220

= 62.5

I = 40A

= 1.5

NDP706AEL

National Semiconductor

NDP603AL

T = 125°C

= 4.5V

T = 25°C

TO-220

TO-263

= 62.5

I = 10A

T = 125°C

= 2.5

National Semiconductor

NDP606AL

= 5V

T = 25°C

= 62.5

I = 24A

T = 125°C

= 1.5

Motorola

= 5V

T = 25°C

= 62.5

I = 37.5A

MTB75N03HDL

Int. Rectifier

T = 125°C

9.3

—

(D PAK)

= 1.0

= 5V

T = 25°C

TO-220

= 62.5

I = 31A

IRLZ44

T = 125°C

= 1.0

Int. Rectifier

= 4.5V

T = 25°C

= 62.5

I = 28A

IRL3103S

T = 125°C

= 1.0

Note:

1. R

) values at Tj = 125°C for most devices were extrapolated from the typical operating curves supplied by the manufac-

DS(ON

turers and are approximations only.

Two MOSFETs in Parallel

We recommend two MOSFETs used in parallel instead of a

single MOSFET. The following signiﬁcant advantages are

realized using two MOSFETs in parallel:

• No added heat sink required.

With the power dissipation down to around one watt and

with MOSFETs mounted ﬂat on the motherboard, no

external heat sink is required. The junction-to-case

thermal resistance for the MOSFET package (TO-220) is

typically at 2°C/W and the motherboard serves as an

excellent heat sink.

• Signiﬁcant reduction of power dissipation.

Maximum current of 14A with one MOSFET:

= (I R

)(Duty Cycle) =

DS,ON

MOSFET

• Higher current capability.

(14) (0.050*)(3.3+0.4)/(5+0.4-0.35) = 7.2 W

With thermal management under control, this on-board

DC-DC converter can deliver load currents up to 14.5A

with no performance or reliability concerns.

With two MOSFETs in parallel:

= (I R

)(Duty Cycle) =

DS,ON

MOSFET

(14/2) (0.037*)(3.3+0.4)/(5+0.4-0.35) = 1.3W/FET

Note: R

increases with temperature. Assume R

= 25mΩ

can easily increase to 50mΩ at high temperature

DS,ON

at 25°C. R

DS,ON

when using a single MOSFET. When using two MOSFETs in

parallel, the temperature effects should not cause the R

to rise

DS,ON

above the listed maximum value of 37mΩ.

AN42

APPLICATION NOTE

12V Gate Bias

MOSFET Gate Bias

Figure 7 illustrates how an external 12V source can be used

to bias VCCQP. A 47 Ω resistor is used to limit the transient

current into the VCCQP pin, and a 1µF capacitor ﬁlter is

used to ﬁlter the VCCQP supply. This method provides a

The MOSFET(s) can be biased using one of two methods:

Charge Pump or 12V Gate Bias.

Charge Pump (or Bootstrap)

higher gate bias voltage (V ) to the MOSFET, and there-

Figure 6 employs a charge pump to provide the MOSFET

gate bias. The charge pump capacitor, CP, is used as a ﬂying

capacitor to boost the voltage of the RC5040 or RC5042 out-

put driver. When the MOSFET switches off, the source of the

MOSFET is at -0.6V. VCCQP is charged through the Schot-

tky diode to 4.5V. Thus, the capacitor CP is charged to 5V.

When the MOSFET turns on, the source of the MOSFET is

at approximately 5V. The capacitor voltage follows, and

hence provides a voltage at VCCQP equal to 10V. The Schot-

tky is required to provide the charge path when the MOSFET

is off, and then reverses bias when the VCCQP goes to 10V.

The capacitor CP needs to be a high Q and high frequency

capacitor. A 1µF ceramic capacitor is recommended here.

fore reduces the R

DS,ON

MOSFET. Figure 8 illustrates how R

and resulting power loss within the

decreases dra-

DS,ON

matically as V increases. A 6.2V Zener (DS2) is used to

clamp the voltage at V

CCQP

to a maximum of 12V and

ensure that the absolute maximum voltage of the IC is not

exceeded.

Warning: The 12V Gate Bias method applies only to the

RC5042. The RC5040 has not been designed to accept an

external 12V gate bias voltage, and may be damaged if

this method is used.

+5V

47Ω

+12V

DS2

6.2V

VCCQP

HIDRV

PWM/PFM

Control

PWM/PFM

Control

DS1

65-AP42-06

65-AP42-07

Figure 6. Charge Pump Configuration

Figure 7. 12V Gate Bias Configuration

0.1

0.09

0.08

0.07

0.06

0.05

0.04

0.03

0.02

0.01

R(DS)Fuji

R(DS)706A

R(DS)-706AEL

1.5

2.5

3.5

Gate-Source Voltage, V

(V)

Figure 8. R

DS,ON

vs. V for Selected MOSFETs

APPLICATION NOTE

AN42

Converter Efﬁciency

Losses due to parasitic resistance in the switches, inductor,

and sense resistor dominate at high load-current levels. The

major loss mechanisms under heavy loads, in order of

importance, are:

• Sense Resistor Losses

• Gate-Charge Losses

• Diode-Conduction Losses

• Transition Losses

• Input Capacitor Losses

• Losses Due to the Operating Supply Current of the IC.

• MOSFET I R Losses

• Inductor Losses

Efﬁciency of the converter under heavy loads can be calculated as follows:

× V

OUT

Efficiency = ------------- = -------------------------------------------------------- ,

× V

+ P

OUT LOSS

OUT

where P

= PD

+ PD

CAP IC

LOSS

MOSFET

INDUCTOR

RSENSE

GATE

DIODE

TRAN

Design Equations:

+ V

OUT

(1) PD

(2) PD

= I

× R

× DutyCycle , whereDutyCycle = -----------------------------------------

DS,ON

MOSFET

OUT

+ V – V

= I

× R

INDUCTOR

OUT

(3) PD

(4) PD

= I

× R

OUT SENSE

RSENSE

= q

× f × 5V , where q

is the gate charge and f is the switching frequency

GATE

(5) PD

= V × I (1 – DutyCycle)

DIODE

× C

× I

× f

LOAD

RSS

(6) PD

(7) PD

= ------------------------------------------------------------- , where C

is the reverse transfer capacitance of the high-side MOSFET.

RSS

TRAN

CAP

DRIVE

= I

× ESR

RMS

(8) PD = V × I

Example:

3.3 + 0.5

DutyCycle = ----------------------------- = 0.73

5 + 0.5 – 0.3

= 10 × 0.030 × 0.73 = 2.19W

MOSFET

= 10 × 0.010 = 1W

INDUCTOR

= 10 × 0.0065 = 0.65W

RSENSE

= CV × f × 5V = 1.75nf × (9 – 1)V × 650Khz × 5V = 0.045W

GATE

= 0.5 × 10(1 – 0.73) = 1.35W

DIODE

5 × 400pf × 10 × 650khz

----------------------------------------------------------------

0.7A

∼ 0.010W

TRAN

= (7.5 – 2.5) × 0.015 = 0.37W

CAP

PD = 0.2W

AN42

APPLICATION NOTE

= 2.19W + 1.0W + 0.65W + 0.045W + 1.35W + 0.010W + 0.37W + 0.2W = 5.815W

LOSS

3.3 × 10

---------------------------------------

∴ Efficiency =

≈ 85%

3.3 × 10 + 5.815

Table 6. RC5040 and RC5042 Short Circuit Comparator

Threshold Voltage

Selecting the Inductor

Selecting the right inductor component is critical in the

DC-DC converter application. The inductor’s critical param-

eters to consider are inductance (L), maximum DC current

Short Circuit Comparator

(mV)

threshold

(I ), and coil resistance (R ).

Typical

Minimum

Maximum

120

100

The inductor core material is crucial in determining the

amount of current it can withstand. As with all engineering

designs, tradeoffs exist between various types of core mate-

rials. In general, Ferrites are popular due to their low cost,

low EMI properties, and high frequency (>500KHz) charac-

teristics. Molypermalloy powder (MPP) materials exhibit

good saturation characteristics, low EMI, and low hysteresis

losses; however, they tend to be expensive and more effec-

tively utilized at operating frequencies below 400KHz.

140

When designing the external current sense circuitry, pay

careful attention to the output limitations during normal

operation and during a fault condition. If the short circuit

protection threshold current is set too low, the converter may

not be able to continuously deliver the maximum CPU load

current. If the threshold level is too high, the output driver

may not be disabled at a safe limit and the resulting power

dissipation within the MOSFET(s) may rise to destructive

levels.

Another critical parameter is the DC winding resistance of

the inductor. This value should typically be as low as possi-

ble because the power loss in DC resistance degrades the

The design equation used to set the short circuit threshold

limit is as follows:

efﬁciency of the converter by P

LOSS

= I x R . The value

of the inductor is a function of the oscillator duty cycle

(T ) and the maximum inductor current (I ). I can be

PK PK

--------

, where: I = output short circuit current

SENSE

calculated from the relationship:

(I – I

)

min

– V

SW D

≥ I

= I

+ ----------------------------

Load, max

inductor

-----------------------------------------

= I

MIN

where I and I

are peak ripple currents and

is the maximum output load current.

pk min

Where T

forward voltage of diode DS1.

is the maximum duty cycle and V is the

load, max

You must also take into account the current (I –I ), or

pk min

The inductor value can be calculated using the following

relationship:

the ripple current ﬂowing through the inductor under normal

operation. Figure 9 illustrates the inductor current waveform

for the RC5040 and RC5042 DC-DC converters at maxi-

mum load.

– V

SW O

-----------------------------------------

L =

– I

MIN

Where V

(R x I ) is the drain-to-source voltage of

SW DS,ON

M1 when it is turned on.

(I – I

)/2

min

Implementing Short Circuit Protection

I_{LOAD, MAX}

min

Intel currently requires all power supply manufacturers

to provide continuous protection against short circuit

conditions that may damage the CPU. To address this

requirement, Raytheon Electronics has implemented a cur-

rent sense methodology on the RC5040 and RC5042 con-

trollers. This methodology limits the power delivered to the

load during an overcurrent condition. The voltage drop cre-

ated by the output current ﬂowing across a sense resistor is

presented to one terminal of an internal comparator with

hysterisis. The other comparator terminal has a threshold

voltage, nominally 120mV. Table 6 states the limits for the

comparator threshold of the switching regulator:

T_ON

T_OFF

T = 1/f

Figure 9. Typical DC-DC Converter

Inductor Current Waveform

The calculation of this ripple current is as follows:

(V –V – V (V + V )

)

(I – I

)

min

OUT

------------------------------------------------ ---------------------------------------------

---------------------------- =

(V – V + V )

IN SW

APPLICATION NOTE

AN42

(I_PK– I_min

)

where:

I_SC≥ I_inductor= I_{Load, max}+ ----------------------------- = 14.5 + 1 = 15.5A

•

= Input Voltage to the Converter

= Voltage Across the MOSFET = I

LOAD

= Forward Voltage of the Schottky Diode

For continuous operation at 14.5A, the short circuit detection

threshold must be at least 15.5A.

x R

DS,ON

T = The Switching Period of the Converter = 1/f^S,

Where f = Switching Frequency.

The next step is to determine the value of the sense resistor.

Including tolerance, the sense resistor value can be approxi-

mated as follows:

For an input voltage of 5V, an output voltage of 3.3V, an

V_th,min

inductor value of 1.3µH, and a switching frequency of

----------------

I_SC

----------------------------------

1.0 + I_Load,max

R_SENSE

× (1 – TF) =

× (1 – TF)

650KHz (using C

EXT

= 39pF), the inductor current can be

calculated as follows:

where TF = Tolerance Factor for the sense resistor.

(I – I

)

min

(5.0 – 14.5 × 0.037 – 3.3)

-------------------------------------------------------------

---------------------------- =

–6

Several different types of sense resistors exist. Table 7

describes tolerance, size, power capability, temperature

coefﬁcient and cost of various sense resistors.

1.3 × 10

(3.3 + 0.5)

-------------------------------------------------------------- -----------------------

= 1.048A

(5.0 – 14.5 × 0.037 + 0.5)

650 × 10

Therefore, for a continued load current of 14.5A, the peak

current through the inductor, I , is found to be:

Table 7. Comparison of Sense Resistors

Discrete Metal

Strip Surface

Mount Resistor

(Dale)

Discrete

CuNi Alloy

Wire Resistor

(Copel)

Discrete Iron

Alloy

Resistor (IRC)

Discrete MnCu

Alloy Wire

Resistor

Motherboard

Trace Resistor

Description

Tolerance

Factor (TF)

±29%

±5%

(±1% available)

±1%

±10%

Size

(L x W x H)

2" x 0.2" x 0.001" 0.45" x 0.065" x 0.25" x 0.125" x 0.200" x 0.04" x

0.200" x 0.04" x

0.100"

(1 oz Cu trace)

0.200"

0.025"

0.160"

Power capability

> 50A/in

1 watt

(3W and 5W

available)

Temperature

Coefficient

+4,000 ppm

+30 ppm

±75 ppm

±30 ppm

±20 ppm

Cost

Low

$0.31

$0.47

$0.09

@10,000 piece

included in

motherboard

Refer to Appendix A for Directory of component suppliers

Based on the Tolerance in the above table, for embedded PC

th,min

----------------------------------------

× (1 – TF) =

trace resistor and for I

= 14.5A:

load,max

SENSE

1.0A + I

Load, max

th,min

----------------------------------------

× (1 – TF) =

SENSE

1.0A + I

Load, max

100mV

---------------------------------

× (1 – 5%) = 6.1mΩ

1.0A + 14.5A

100mV

---------------------------------

1.0A + 14.5A

× (1 – 29%) = 4.6mΩ

For user convenience, Table 8 lists the recommended values

for sense resistor at various load currents using an embedded

PC trace resistor or discrete resistor.

For a discrete resistor and I

= 14.5A:

load, max

AN42

APPLICATION NOTE

Table 8. R

for various load currents

Table 9 is a summary of tolerances for the Embedded PC

Trace Resistor.

sense

SENSE

PC Trace

Resistor (mΩ)

Discrete

Resistor (mΩ)

Load,max

Table 9. Summary PC Trace Resistor Tolerance

(A)

Tolerance due to sheet resistivity variation

Tolerance due to L/W error

16%

10.0

11.2

12.4

13.9

14.0

14.5

6.5

5.8

5.3

4.8

4.7

4.6

8.6

7.8

7.1

6.4

6.3

6.1

Tolerance due to temperature variation

Total Tolerance for PC Trace Resistor

12%

29%

Design rules for using an embedded resistor

The basic equation for laying an embedded resistor is:

W × t

-------------

R = ρ ×

Discrete Sense Resistor

Discrete iron alloy resistors come in a variety of tolerances

and power ratings, and are ideal for precision implementa-

tions. Either an MnCu alloy wire resistor or an CuNi alloy

wire resistor is ideal for a low cost implementation.

where ρ is the Resistivity (W-mil), L is the Length (mils), W

is the Width (mils), and t is the Thickness (mils).

For 1oz copper, t = 1.35 mils, ρ = 717.86 µΩ-mil,

1 L/1 W = 1 Square ( ■ ).

For example, you can layout a 5.30mΩ embedded sense

Embedded Sense Resistor (PC Trace Resistor)

resistor. From Equations above,

Embedded PC trace resistors have the advantage of almost

zero cost implementation. However, the value of the PC

trace resistors have large variations. Embedded resistors

have 3 major error sources: the sheet resistivity of the inner

layer, the mismatch due to L/W, and the temperature varia-

tion of the resistor. When laying out embedded sense resis-

tors, consider all error sources described as follows:

0.05

W = --------- = --------- = 200mil

0.05

R × W × t

0.00530 × 200 × 1.35

L = ----------------------- = --------------------------------------------------- = 2000mi

717.86

L/W = 10 ■.

Therefore, to model 5.30mΩ enbedded resistor, you need

• Sheet resistivity.

W = 200 mils, and L = 2000 mils. See Figure 10.

For 1 ounce copper, the thickness variation is typically

between 1.15 mil and 1.35 mil. Therefore, the error due to

sheet resistivity is (1.35 – 1.15)/1.25 = 16%.

W = 200 mils

• Mismatch due to L/W.

The error in L/W is dictated by the geometry and the

power dissipation capability of the sense resistor. The

sense resistor must be able to handle the load current and,

therefore, requires a minimum width, calculated as

follows:

L = 2000

Figure 10. 5.30mΩ Sense Resistor (10 ■)

You can also implement the sense resistor in the following

manner. Each corner square is counted as 0.6 square since

the current ﬂowing through the corner square does not ﬂow

uniformly, concentrated towards the inside edge. This is

shown in Figure 11.

W = ---------

0.05

where W is the minimum width required for proper power

dissipation (mils), and I is the load current in Amps.

For a load current of 15A, the minimum width required is

300mils, which reﬂects a 1% L/W error.

• Thermal Considerations.

The I R power losses cause the surface temperature of the

Figure 11. 5.30mΩ Sense Resistor (10 ■)

resistor to increase along with its resistance value. In

addition, ambient temperature variations add the change

in resistor value:

A Resign Example Combining an Embedded Resistor

with a Discrete Resistor

R = R [1 + α (T – 20)]

For low cost implementation, the embedded PC trace resistor

is the most desirable alternative, but, as discussed earlier, the

wide tolerance (±29%) presents a challenge. In addition,

changing CPU requirements may force the maximum load

where R is the resistance at 20°C, α = 0.00393/ °C,T

is the operating temperature, andR is the desired value.

For temperature T = 50°C, the %R change = 12%.

APPLICATION NOTE

AN42

Embedded Sense Resistor

IFBH

MnCu Discrete

Resistor

R21

R22

IFBL

Output Power

Plane (Vout)

R-∆r

R+∆r

Figure 12. Short Circuit Sense Resistor Design Using PC Trace Resistor and Optional Discrete Sense Resistor

currents to change. Therefore, combining an embedded

resistor with a discrete resistor may be a desirable option.

This section discusses a design that provides ﬂexibility and

addresses wide tolerances. Refer to Figure 12.

The converter exhibits at normal load regulation until the

voltage across the resistor reaches the internal short circuit

threshold of 120mV. At this point, the internal

comparator trips and signals the controller to turn off the

gate drive to the power MOSFET. This causes a drastic

reduction in the output voltage as the load regulation col-

lapses into the short circuit control mode. The output voltage

does not return to its nominal value until the output short cir-

cuit current is reduced to within the safe range for the DC-

DC converter.

In this design, the user has the option to choose either an

embedded or a discrete MnCu sense resistor. To use the dis-

crete sense resistor, populate R21 with a shorting bar (zero

Ohm resistor) for a proper Kelvin connection and add the

MnCu sense resistor. To use the embedded sense resistor,

populate R22 with a shorting bar for a Kelvin connection.

The embedded sense resistor allows you to choose a plus or a

minus delta resistance tap to offset any large sheet resistivity

change.

Power Dissipation Consideration During a

Short Circuit Condition

The RC5040 and RC5042 controllers respond to an output

short circuit by drastically changing the duty cycle of the

gate drive signal to the power MOSFET. In doing this, the

power MOSFET is protected from over-stress and eventual

destruction. Figure 14A shows the gate drive signal of a typ-

ical RC5040 operating in continuous mode with a load cur-

rent of 10A. The duty cycle is then set by the ratio of the

input voltage to the output voltage. If the input voltage is 5V

and the output voltage is 3.1V, the ratio of Vout/ Vin is 62%.

Figure 14B shows the result of the RC5040 going into its

short circuit mode when the duty cycle is around 20%. Cal-

culating the power on the MOSFET at each condition on the

graph in Figure 13 shows how the protection scheme works.

The power dissipated in the MOSFET at normal operation

for a load current of 14.5A, is given by:

In this design, the center tap yields 6mΩ, and the left or the

right tap yield 6.7 or 5.3 mΩ, respectively.

RC5040 and RC5042 Short Circuit Current

Characteristics

The RC5040 and RC5042 have a short circuit current char-

acteristic that includes a hysteresis function. This function

prevents the DC-DC converter from oscillating in the event

of a short circuit. Figure 13 shows the typical characteristic

of the DC-DC converter using a 6.5 mΩ sense resistor.

3.5

²× .037 × .62 = 1.2W

3.0

2.5

2.0

1.5

1.0

14.5

P_D= I²× R_ON× DutyCycle =

---------

for each MOSFET.

The power dissipated in the MOSFET at short circuit

condition for a peak short current of 20A, is given by:

0.5

-----

× .037 × .2 = 0.74W

for each MOSFET.

Output Current

Figure 13. RC5040/RC5042 Short Circuit Characteristic

Thus, the MOSFET is not being over-stressed during a short

circuit condition.

AN42

APPLICATION NOTE

= I

× V × (1 – DutyCycle) =

F, ave

D, Diode

14 × 0.45 × 0.8 ≈ 5W

Thus for the Schottky diode, the thermal dissipation during

a short circuit is greatly magniﬁed and requires that the

thermal dissipation of the diode be properly managed by the

appropriate choice of a heat sink. In order to protect the

Schottky from being destroyed in the event of a short, we

should limit the junction temperature to less than 130°C.

Using the equation for maximum junction temperature,

we can arrive at the thermal resistance required below:

– T

J(max)

= -------------------------------

ΘJA

Figure 14A. V

CCQP

Output Waveform for Normal

= 3.3V@10A

Operation Condition with V

Assuming that the ambient temperature is 50°C, we get:

– T

out

130 – 50

J(max)

= ------------------------------- = -------------------- = 16°C ⁄ W

ΘJA

Thus we need to provide for a heat sink that will give the

Schottky diode a thermal resistance of at least 16°C/W or

lower in order to protect the device during an indeﬁnite

short.

In summary, with proper heat sink, the Schottky diode is not

being over stressed during a short circuit condition.

Schottky Diode Selection

The application circuits of Figures 3, 4, and 5 show two

Schottky diodes, DS1 and DS2. In synchronous mode, DS1

is used in parallel with M3 to prevent the lossy diode in the

FET from turning on. In non-synchronous mode, DS1 is

used as a ﬂyback diode to provide a constant current path for

the inductor when M1 is turned off.

Figure 14B. V

CCQP

Output Waveform for

Output Shorted to Ground

The Schottky diode has a power dissipation consideration

during the short circuit condition. During normal operation,

the diode dissipates power when the power MOSFET is off.

The power dissipation is given by:

The Schottky diode DS2 serves a dual purpose. As conﬁg-

ured in Figures 3, 4, and 5, DS2 allows the VCCQP pin on

the RC5040 to be bootstrapped up to 9V using capacitor

C12. When the lower MOSFET M3 is turned on, one side of

capacitor C12 is connected to ground while the other side of

the capacitor is being charged up to voltage VIN – VD

through DS2. The voltage that is then applied to the gate of

the MOSFET is VCCQP – VSAT, or typically around 9V.

DS2 also provides correct sequencing of the various supply

voltages by assuring that VCCQP is not enabled before the

other supplies.

= I × V × (1 – DutyCycle) =

D, Diode

14.5 × 0.5V × (1 – 0.62) = 2.75W

In short circuit mode, the duty cycle is dramatically reduced

to approximately 20%. The forward current during a short

circuit condition decays exponentially through the inductor.

The power dissipated on the diode during the short circuit

condition, is approximated by:

A vital selection criteria for DS1 and DS2 is that they exhibit

a very low forward voltage drop, as this parameter can

directly affect the regulator efﬁciency. Table 10 lists several

suitable Schottky diodes. Note that the MBR2015CTL has a

very low forward voltage drop. This diode is ideal for appli-

cations where output voltages less than 2.8V are required.

1.5us

– -------------

1.3us

– -----------

L ⁄ R

= I × e

= 20A × e

≈ 7.9A

F, ending

≈ (20A + 7.9A) ⁄ 2 ≈ 14A

F, ave

APPLICATION NOTE

AN42

With this in mind, correct calculation of the output capaci-

Table 10. Schottky Diode Selection Table

tance is crucial to the performance of the DC-DC converter.

The output capacitor determines the overall loop stability,

output voltage ripple, and load transient response. The calcu-

lation is as follows:

Manufacturer

Model #

Forward Voltage

Conditions

V_F

Philips

PBYR1035

I_F= 20A; T_j=25°C

I_F= 20A; T_j=125°C

< 0.84v

< 0.72v

I × ∆T

Motorola

MBR2035CT I_F= 20A; T_j=125°C

I_F= 20A; T_j=25°C

< 0.84v

< 0.72v

C(µF) = -------------------------------------

∆V – I × ESR

Motorola

MBR1545CT I_F= 15A; T_j=125°C

I_F= 15A; T_j=25°C

< 0.84v

< 0.72v

where ∆V is the maximum voltage deviation due to load

transients, ∆T is the reaction time of the power source, and

is the output load current. ∆V is the loop response time of

Motorola

MBR2015CTL I_F= 20A; T_j=150°C

I_F= 20A; T_j=25°C

< 0.58v

< 0.48v

the RC5040 and RC5042, approximately 8µs.

For I = 10A and ∆V = 165mV, the bulk capacitance

Output Filter Capacitors

required can be approximated as follows:

Output ripple performance and transient response are

functions of the ﬁlter capacitors. Since the 5V supply of a PC

motherboard may be located several inches away from the

DC-DC converter, the input capacitance can play an impor-

tant role in the load transient response of the RC5040.

The higher the input capacitance, the more charge storage is

available for improving the current transfer through the

FET(s). Capacitors with low Equivalent Series Resistance

(ESR) are best for this type of application and can inﬂuence

the converter's efﬁciency if not chosen carefully. The input

capacitor should be placed as close to the drain of the FET as

possible to reduce the effect of ringing caused by long trace

lengths.

I_O× ∆T

C(µF) = ------------------------------------- = --------------------------------------------------------- = 1454µF

∆V – I_O× ESR 165mV – 10A × 11mΩ

10A × 8µs

Input ﬁlter

The DC-DC converter design should include an input induc-

tor between the system +5V supply and the converter input

as described below. This inductor will serve to isolate the

+5V supply from noise occurring in the switching portion of

the DC-DC converter and also to limit the inrush current into

the input capacitors during power up. An inductor value of

around 2.5µH is recommended, as illustrated in Figure 15.

2.5µH

Vin

ESR is the resonant impedance of the capacitor, and it is dif-

ﬁcult to quantify. Since the capacitor is actually a complex

impedance device having resistance, inductance, and capaci-

tance, it is natural for it to have a resonant frequency. As a

rule, the lower the ESR, the better suited the capacitor is for

use in switching power supply applications. Many manufac-

turers do not supply ESR data, but a useful estimate can be

obtained using the following equation:

1000µF, 10V

0.1µF

Electrolytic

65-AP42-17

Figure 15. Input Filter

ESR = ------------

2πfC

Bill of Materials

where DF is the dissipation factor of the capacitor, f is the

operating frequency, and C is the capacitance in farads.

The Bill of Materials for the application circuits of Figures 2

through 4 is presented in Table 11.

Table 11. Bill of Materials for a 14.5A Pentium Pro Motherboard Application

C4, C5, C7, C8, C9,

C10

Panasonic

ECU-V1H104ZFX

0.1µF 50V capacitor

4.7µF 16V capacitor

39pF capacitor

Panasonic

ECSH1CY475R

Cext

Panasonic

ECU-V1H121JCG

C12

C1, C2, C3

United Chemicon

LXF16VB102M

1000µF 6.3V electrolytic

capacitor 10mm x 20mm

ESR<0.047Ω

C11

Panasonic

ECU-V1H224ZFX

0.22µF 50V capacitor

AN42

APPLICATION NOTE

Table 11. Bill of Materials for a 14.5A Pentium Pro Motherboard Application

C13, C14, C15

Sanyo

6MV1500GX

1500µF 6.3V electrolytic

ESR < 0.047 Ω

Vf<0.72V @ If = 15A

1A, 20V

capacitor 10mm x 20mm

DS1

(note 1)

Motorola

MBR1545CT

Shottky Diode

DS2

General Instruments 1N5817

Skynet 320-8107

Schottky Diode

1.3µH inductor

2.5µH inductor

L2*

Skynet

320-6110

*Optional – will help re-

duce ripple on 5v line

M1, M2, M3

(note 2)

Fuji

2SK1388

N-Channel Logic Level

Enhancement Mode MOSFET

< 37m ohm

DS(ON)

< 4V, I > 20A

Rsense

COPEL

A.W.G. #18

6 milliohm CuNi Alloy Wire

resistor

R1, R2, R3, R4, R6,

Panasonic ERJ-6ENF10.0KV

10K 5% Resistors

Raytheon

RC5042M or RC5040M

DC-DC Converter for Pentium

Pro

Refer to Appendix A for Directory of component suppliers.

Notes:

1. In synchronous mode using the RC5040, a 1A schottky diode (1N5817) may be substituted for the MBR1545CT.

2. MOSFET M3 is only required for the RC5040 synchronous application.

PCB Layout Guidelines and Considerations

PCB Layout Guidelines

• The CEXT timing capacitor should be surrounded with a

• Placement of the MOSFETs relative to the RC5040 is

ground trace. The placement of a ground or power plane

critical. The MOSFETs (M1 & M2), should be placed

underneath the capacitor provides further noise isolation,

such that the trace length of the HIDRV pin to the FET

and helps to shield the oscillator from the noise on the

gate is minimized. A long lead length causes high

PCB. This capacitor should be placed as close to pin 1 as

amounts of ringing due to the inductance of the trace and

possible.

the large gate capacitance of the FET. This noise radiates

all over the board, and because it is switching at a high

• Group the MOSFETs, inductor, and Schottky diode as

voltage and frequency, it is very difﬁcult to suppress.

close together as possible. This minimizes ringing derived

from the inductance of the trace and the large gate

Figure 16 shows an example of proper MOSFET

capacitance of the FET. Place the input bulk capacitors as

placement in relation to the RC5040. It also shows an

close to the drains of MOSFETs as possible. In addition,

example of problematic placement for the MOSFETs.

place the 0.1µF decoupling capacitors right on the drain

of each MOSFET. This helps to suppress some of the high

In general, noisy switching lines should be kept away

frequency switching noise on the DC-DC converter input.

from the quiet analog section of the RC5040. That is,

traces that connect to pins 12 and 13 (HIDRV and

• The traces that run from the RC5040 IFB (pin 4) and VFB

VCCQP) should be kept far away from the traces that

(pin 5) pins should be run next to each other and be Kelvin

connect to pins 1 through 5, and pin 16.

connected to the sense resistor. Running these lines

together helps to reject some of the common mode noise

to the RC5040 feedback input. Run the noisy switching

• Place the 0.1µF decoupling capacitors as close to the

RC5040 and RC5042 pins as possible. Extra lead length

signals (HIDRV & VCCQP) on one layer, and use the

negates their ability to suppress noise.

inner layers for power and ground only. If the top layer is

being used to route all of the noisy switching signals, use

• Each VCC and GND pin should have its own via to the

the bottom layer to route the analog sensing signals VFB

appropriate plane on the board to add isolation between

and IFB.

pins

APPLICATION NOTE

AN42

Good layout

Bad layout

11 RC5040

= “Quiet” Pins

Figure 16. Example of Proper MOSFETs Placements

File can be obtained from Raytheon Electronics Semicon-

ductor Division’s Marketing Department at (415) 966-7819.

PC Motherboard Layout and Gerber File

A reference design for motherboard implementation of the

RC5040 and RC5042 along with the Layout Gerber File and

Silk Screen are presented below. The actual PCAD Gerber

AN42

APPLICATION NOTE

AN42

5. Apply load at 1A increments; an active load (HP6060B

or equivalent) is suggested.

Guidelines for Debugging and

Performance Evaluations

6. In case of poor regulation, refer to the procedures in the

DebuggingYour First Design Implementation

Troubleshooting section.

Use the following procedure to help you debug your design

implementation:

Troubleshooting

1. If no voltage is registered at the output and the circuit is

not drawing current, look for openings in the connec-

tions. Check the circuitry versus the schematic, and the

power supply pins at the device to ascertain that volt-

age(s) had been applied.

1. Note the VID pins settings. They tell you what voltage is

to be expected.

2. Do not connect any load to the circuit. While monitoring

the output voltage, apply power to the part with current

limiting at the power supply. Do this to make sure that

no catastrophic shorts occur.

2. If no voltage is registered at the output and the circuit is

drawing excessive current (>100mA) with no load,

check for possible shorts. Trace the path of the excessive

current to determine if the controller is at fault or if the

excessive current is due to peripheral components.

3. Ιf proper voltage is not achieved, follow the procedures

in the Troubleshooting section.

3. If the output voltage comes near to, but is not, what is

expected, check the VID inputs at the device pins. The

part is factory set to correspond to the VID inputs.

4. After there is proper voltage, increase the current limit-

ing of the power supply to 16A.

AN42

APPLICATION NOTE

Load Regulation

4. Premature shut down can be caused by an inappropriate

value of sense resistor. See the Sense Resistor section.

VID

I_load(A)

0.5

V_out(V)

3.0904

3.0825

3.0786

3.0730

3.0695

3.0693

3.0695

3.0694

3.0691

0.70%

5. A poor load regulation can have many causes. You

should ﬁrst check the voltages and signals at the critical

pins.

0100

1.0

2.0

6. The VREF pin should be at the voltage set by the VID

pins. If the power supply pins are correct and the VID

pins are correct, the VREF should be at the correct volt-

age.

3.0

4.0

5.0

7. Next check the oscillator pin. A saw tooth wave at the

frequency set by the external capacitor should be seen.

6.0

7.0

8. When the VREF and CEXT pins are determined to be

correct and the output voltage is still incorrect look at

the waveform at VCCQP. This pin should be swinging

from ground to +12V (in the +12V application) and

from slightly below +5V to about +10V (charge pump

application). If the VCCQP pin is noisy, with ripples and

overshoots, then the noise may cause the converter to

function improperly.

8.0

9.0

9.9

Load Regulation 0.5A – 9.9A

VID

I_load(A)

0.5

V_out(V)

3.2805

3.2741

3.2701

3.2642

3.2595

3.2597

3.2606

3.2611

3.2613

3.2611

3.2607

3.2599

3.2596

0.64%

0010

9. Next, look at the HIDRV pin. This pin directly drives the

gate of the FET. It should provide a gate drive (Vgs) of

about 5V when turning the FET on. A careful study of

the layout is recommended. See the PCB Layout Guide-

lines and Considerations section.

1.0

2.0

3.0

4.0

10. Experience shows that the most frequent errors are using

incorrect components, improper connections, and poor

layout.

5.0

6.0

7.0

Performance Evaluation

8.0

This section shows the results of a random sample evalua-

tion. Use these results as a reference guide for evaluating the

RC5040 DC-DC converter for Pentium Pro motherboards.

9.0

10.0

11.0

12.0

12.4

Load Regulation 0.5A – 12.4A

APPLICATION NOTE

AN42

VID

(A)

(V)

Low to High 0.5A-9.9A

Current Step

- 76.0mV

+ 70mV

Refer to

load

out

Attachment

A for Scope

Picture

1010

0.5

2.505

2.504

2.501

2.496

2.493

2.492

2.491

2.490

2.489

2.488

2.486

2.485

2.484

0.84%

1.0

2.0

High to Low 9.9A-0.5A

Current Step

Refer to

Attachment

B for Scope

Picture

3.0

4.0

5.0

Low to High 0.5A-12.4A - 97.6mV

Current Step

Refer to

6.0

Attachment

C for Scope

Picture

7.0

8.0

High to Low 12.4A-0.5A + 80.0mV

Current Step

Refer to

9.0

Attachment

D for Scope

Picture

10.0

11.0

12.0

13.0

13.9

Low to High 0.5A-13.9A - 99.2mV

Current Step

Refer to

Attachment

E for Scope

Picture

High to Low 13.9A-0.5A + 105.2mV Refer to

Current Step

Load Regulation 0.5 - 13.9A

Attachment

Note:

F for Scope

Picture

Load regulation is expected to be typically around 0.8%. The

load regulation performance for this device under evaluation is

excellent.

Note:

Excellent transient voltage response. Transient voltage is rec-

ommended to be less than 4% of the output voltage. The per-

formance of the device under evaluation is significantly better

than a typical VRM.

Output Voltage LoadTransients Due to Load Current Step

This test is performed using Intel P6.0/P6S/P6T Voltage

Transient Tester.

Input Ripple and Power on Input Rush Current

I_load= 9.9A Input Ripple

Refer to Attachment

Voltage = 15mV G for Scope Picture

Power on Input Rush Current is not measured on the mother-

board because we did not want to cut the 5V trace and insert

current probe in series with the supply. However, with the

input ﬁlter design, the Input Rush Current will be well within

speciﬁcation.

AN42

APPLICATION NOTE

Component Case Temperature

Case Temperature Case Temperature Case Temperature

(°C)

I_load= 9.9A

(°C)

I_load= 12.4A

(°C)

I_load=13.9A

Device

Description

Q3A

MOSFET

K1388

56.3

66.6

61.2

Q3B

MOSFET

K1388

Inductor,

Unknown

Schottky Diode

2048CT

Raytheon RC5040

38.2

Cin

Cout

Input Capacitor 1000µF

36.8

34.8

Output Capacitor

38.2

1500µF

Note:

Case temperatures are all within guidelines. Our guideline is that case temperatures for all components should be below 105°C

@25°C Ambient.

Comments:

Excellent input ripple voltage. Input ripple voltage is recommended to be less than 5% of the output voltage.

Evaluation Summary:

The on-board DC-DC converter is fully functional. It has

excellent load regulation, transient response, and input

voltage ripple.

Attachment A

Attachment B

APPLICATION NOTE

AN42

Attachment E

Attachment C

Attachment F

Attachment D

Attachment G

Summary

RC5040/RC5042 Evaluation Board

This application note covers for implementation of a DC-DC

converter on a Pentium Pro motherboard using the RC5040

and RC5042. The detailed discussion includes Pentium Pro

processor power requirements, RC5040 and RC5042

description, design considerationsn and component selec-

tions, layout guidelines and considerations, guidelines for

debugging, and performance evaluations.

Raytheon Electronics provides an evaluation board for the

purpose of verifying system level performance of the

RC5040 and RC5042. The evaluation board serves as a guide

as to what can be expected in performance with the supplied

external components and PCB layout. Please call Raytheon

Electronics Marketing Department at (415) 966-7819 for an

evaluation board.

AN42

APPLICATION NOTE

Appendix A: Directory of Component Suppliers

Dale Electronics, Inc.

E. Hwy. 50, PO Box 180

Yankton, SD 57078-0180

PH: (605) 665-9301

National Semiconductor

2900 Semiconductor Drive

Santa Clara, CA 95052-8090

PH: (800) 272-9959

Fuji Electric

Nihon Inter Electronics Corp.

Quantum Marketing Int’l, Inc.

12900 Rolling Oaks Rd.

Caliente, CA 93518

Collmer Semiconductor Inc.

14368 Proton Rd.

Dallas, Texas 75244

PH: (214)233-1589

PH: (805) 867-2555

General Instrument

Panasonic Industrial Co.

6550 Katella Avenue

Cypress, CA 90630

PH: (714) 373-7366

Power Semiconductor Division

10 Melville Park Road

Melville, NY 11747

PH: (516) 847-3000

Pulse Engineering

Hoskins Manufacturing Co.

(Copel Resistor Wire)

10776 Hall Road

12220 World Trade Drive

San Diego, CA 92128

PH: (619) 674-8100

Hamburg, MI 48139-0218

PH: (313) 231-1900

Sanyo Energy USA

2001 Sanyo Avenue

San Diego, CA 92173

PH: (619) 661-6620

Intel Corp.

5200 NE Elam Young Pkwy.

Hillsboro, OR. 97123

PH: (800) 843-4481 Tech. Support

for Power Validator

Siliconix

Temic Semiconductors

2201 Laurelwood Road

Santa Clara, CA 95056-1595

PH: (800) 554-5565

International Rectifier

233 Kansas St.

El Segundo, CA 90245

PH: (310) 322-3331

Sumida Electric USA

5999 New Wilke Road Suite #110

Rolling Meadows, IL 60008

PH: (708) 956-0702

IRC Inc.

PO Box 1860

Boone, NC 28607

PH: (704) 264-8861

Xicon Capacitors

PO Box 170537

Motorola Semiconductors

PO Box 20912

Arlington, Texas 76003

PH:(800) 628-0544

Phoenix, Arizona 85036

PH:(602) 897-5056

LIFE SUPPORT POLICY

FAIRCHILD’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES

OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR

CORPORATION. As used herein:

1. Life support devices or systems are devices or systems

which, (a) are intended for surgical implant into the body,

or (b) support or sustain life, and (c) whose failure to

perform when properly used in accordance with

instructions for use provided in the labeling, can be

reasonable expected to result in a significant injury of the

user.

2. A critical component in any component of a life support

device or system whose failure to perform can be

reasonably expected to cause the failure of the life support

device or system, or to affect its safety or effectiveness.

Fairchild Semiconductor

Corporation

Fairchild Semiconductor

Europe

Fairchild Semiconductor

Hong Kong Ltd.

National Semiconductor

Japan Ltd.

Americas

Fax: +49 (0) 1 80-530 85 86

13th Floor, Straight Block,

Ocean Center, 5 Canto Rd.

Tsimshatsui, Kowloon

Hong Kong

Tel:81-3-5620-6175

Fax:81-3-5620-6179

Customer Response Center

Tel:1-888-522-5372

Email: [email protected]

Deutsch Tel: +49 (0) 8 141-35-0

English Tel: +44 (0) 1 793-85-68-56

Italy

Tel: +39 (0) 2 57 5631

Tel:+852 2737-7200

Fax:+852 2314-0061

www.fairchildsemi.com

2/98 0.0m

Stock#AN30000042

1998 Fairchild Semiconductor Corporation

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