Fairchild SEMICONDUCTOR RC5042 User Manual

Application Note 42  
Implementing the RC5040 and RC5042  
DC-DC Converters on Pentium® Pro Motherboards  
Introduction  
PentiumPro and OverDrive®  
Processor Power Requirements  
This document describes how to implement a switching volt-  
age regulator using an RC5040 or an RC5042 high speed  
controller, a power inductor, a Schottky diode, appropriate  
capacitors, and external power MOSFETs. This regulator  
forms a step down DC-DC converter that can deliver up to  
14.5A of continuous load current at voltages ranging from  
2.1V to 3.5V. A specific application circuit, design consider-  
ations, component selection, PCB layout guidelines and per-  
formance evaluation procedures are covered in detail.  
Use Intel’s AP-523 Application Note, Pentium® Pro  
Processor Power Distribution Guidelines, November 1995  
(order number 242764-001), as a basic reference. The speci-  
fications contained in this document have been modified  
slightly from the original Intel document to include updated  
specifications for Pentium Pro microprocessors. Please con-  
tact Intel Corporation for specific details.  
Input Voltages  
In the past 10 years, microprocessors have evolved at such an  
exponential rate that a modern chip can rival the computing  
power of a mainframe computer. Such evolution has been  
possible because of the increasing numbers of transistors that  
processors integrate. Pentium CPUs, for example, integrate  
well over 5 million transistors on a single piece of silicon.  
Available inputs are +5V ±5% and +12V ±5%. Raytheon  
Electronics’ DC-DC converters may use either or both  
inputs. Their input voltage requirements are listed in Table 1.  
Table 1. Input Voltage Requirements  
Controller  
MOSFET  
Drain  
MOSFET  
Gate Bias  
To integrate so many transistors on a piece of silicon, their  
physical geometry has been reduced to the sub-micron level.  
As a result of each geometry reduction, the corresponding  
operational voltage for each transistor has also been reduced.  
This changing voltage for the CPU demands the design of a  
programmable power supply—a design that is not com-  
pletely re-engineered with every change in CPU voltage.  
Part #  
V
CC  
RC5040  
RC5042  
+5V ±5%  
+5V ±5%  
+5V ±5% or  
12V ±5%  
RC5043  
+5V ±5%  
12V ±5%  
12V ±5%  
Pentium Pro DC Power Requirements  
Refer to Table 2 for the power supply specifications for  
Pentium Pro and Overdrive Processors. For a motherboard  
design without a standard Voltage Regulator Module (VRM)  
socket, the on-board DC-DC converter must supply a mini-  
mum I P current of 13.9A at 2.5V and 12.4A at 3.3V. For a  
CC  
flexible motherboard design, the on-board converter must be  
The operational voltage of CPUs has shown a downwards  
trend for the past 5 years: from 5V for the x386 and x486, to  
3.3V for Pentium, and 3.1V for Pentium Pro. Furthermore,  
emerging chip technologies may require operating voltages  
as low as 2.5V. With this trend in mind, Raytheon Electron-  
ics has designed the RC5040 and RC5042 controllers. These  
controllers integrate the necessary programmability to  
address the changing power supply requirements of lower  
voltage CPUs.  
able to supply 14.5A maximum I P.  
CC  
DC Voltage Regulation  
As indicated in Table 2, the voltage level supplied to the  
CPU must be within ±5% of its nominal setting. Voltage  
regulation limits must include:  
Previous generations of DC-DC converter controllers were  
designed with fixed output voltages adjustable only with a  
set of external resistors. In a high volume production envi-  
ronment (such as with personal computers), however, a CPU  
voltage change requires a CPU board re-design to accommo-  
date the new voltage requirement. The integrated 4-bit DAC  
in the RC5040 and the RC5042 reads the voltage ID code  
from the Pentium Pro microprocessor and configures the sys-  
tem to provide the appropriate voltage. In this manner, the  
PC board does not have to be re-designed each time the CPU  
voltage changes. The CPU can thus automatically configure  
its own required voltage.  
• Output load ranges specified in Table 2  
• Output ripple/noise  
• DC output initial voltage set point  
• Temperature and warm up drift (Ambient +10°C to +60°C  
at full load with a maximum rate of change of 5°C per 10  
minutes minimum but no more than 10°C per hour)  
• Output load transient with:  
Slew rate >30A/µs at the converter pins  
Range: 0.3A – I P Max (as defined in Table 2).  
CC  
Rev. 1.1.0  
APPLICATION NOTE  
AN42  
The RC5040 and RC5042 Controllers  
RC5040 and RC5042 Description  
The RC5040 is a programmable synchronous-mode DC-DC  
converter controller. The RC5042 is a non-synchronous ver-  
sion of the RC5040. When designed with the appropriate  
external components, either device can be configured to  
deliver more than 14.5A of output current. During heavy  
loading conditions, these controllers function as current-  
mode PWM step-down regulators. Under light loads, they  
function in PFM (pulse frequency modulation) or pulse skip-  
ping mode. The controllers sense the load level and switch  
between the two operating modes automatically, thus opti-  
mizing efficiency under all loads. The key differences  
between the RC5040 and RC5042 are listed in Table 4.  
Simple Step-Down Converter  
S1  
L1  
+
VIN  
D1  
C1  
RL Vout  
65-AP42-01  
Figure 1. Simple Buck DC-DC Converter  
Figure 1 illustrates a step-down DC-DC converter with no  
feedback control. The basic step-down converter serves as  
the basis for deriving the design equations for the RC5040  
and RC5042. From Figure 1, the basic operation begins by  
closing the switch S1, so that the input voltage V is  
IN  
impressed across inductor L1. The current flowing through  
this inductor is given by the following equation:  
Table 4. RC5040 and RC5042 Differences  
RC5040  
RC5042  
Operation  
Package  
Synchronous Non-Synchronous  
20-pin SOIC  
Yes  
16-pin SOIC  
No  
Output Enable/  
Disable  
(V – V  
)T  
OUT ON  
IN  
I
= -----------------------------------------------  
L
L1  
Refer to the RC5040 Block Diagram illustrated in Figure 2.  
The control loop of the regulator contains two main sections:  
the analog control block and the digital control block. The  
analog block consists of signal conditioning amplifiers feed-  
ing into a set of comparators which provide the inputs to the  
digital block. The signal conditioning section accepts inputs  
from the IFB (current feedback) and VFB (voltage feedback)  
pins and sets two controlling signal paths. The voltage con-  
trol path amplifies the VFB signal and presents the output to  
one of the summing amplifier inputs. The current control  
path takes the difference between the IFB and VFB and pre-  
sents the result to another input of the summing amplifier.  
These two signals are then summed together with the slope  
compensation input from the oscillator. This output is then  
presented to a comparator, which provides the main PWM  
control signal to the digital control block.  
where T is the duty cycle (the time when S1 is closed).  
ON  
When S1 opens, the diode D1 conducts the inductor  
current and the output current is delivered to the load accord-  
ing to the following equation:  
V
(T – T  
L1  
)
ON  
OUT  
S
I
= -------------------------------------------  
L
where T is the overall switching period and (T – T ) is  
S
S
ON  
the time during which S1 is open.  
By solving these equations you can obtain the basic relation-  
ship for the output voltage of a step-down converter:  
T
ON  
----------  
V
= V  
IN  
OUT  
T
The additional comparators in the analog control section sets  
the threshold for when the RC5040 enters PFM mode during  
light loads and the point when the current limit comparator  
disables the output drive signals to the MOSFETs.  
S
In order to obtain a more accurate approximation for V  
,
OUT  
we must also include the forward voltage V across diode  
D
D1 and the switching loss, V . After taking into account  
SW  
these factors, the new relationship becomes:  
The digital control block is designed to take the comparator  
inputs along with the main clock signal from the oscillator  
and provide the appropriate pulses to the HIDRV and  
LODRV pins that control the external power MOSFETs. The  
digital section was designed utilizing high speed Schottky  
transistor logic, thus allowing the RC5040 to operate at clock  
speeds as high as 1MHz.  
T
ON  
----------  
V
= (V + V – V  
)
– V  
D
OUT  
IN  
D
SW  
T
S
Where V  
SW  
= I • R .  
DS,ON  
L
3
AN42  
APPLICATION NOTE  
Main Control Loop  
RC5040  
+5V  
VIN  
OSCILLATOR  
+
+
+
VO  
+
DIGITAL  
CONTROL  
1.24V  
REFERENCE  
4-BIT  
DAC  
VREF  
POWER  
GOOD  
PWRGD  
65-5040-01  
VID0  
VID1  
VID2  
VID3  
Figure 2. RC5040 Block Diagram  
High Current Output Drivers  
Power Good (PWRGD)  
The RC5040 contains two identical high current output  
drivers that use high speed bipolar transistors in a push-pull  
configuration. Each driver is capable of delivering 1A of cur-  
rent in less than 100ns. Each driver’s power and ground are  
separated from the chip power and ground for additional  
switching noise immunity. The HIDRV driver’s power sup-  
ply, VCCQP, is boot-strapped from a flying capacitor as  
illustrated in Figure 3. Using this configuration, C12 is  
charged from VCC via the Schottky diode DS2 and boosted  
when the FET is turned on. This scheme provides a VCCQP  
voltage equal to 2•VCC – VDS(DS2), or approximately 9.5V  
when VCC = 5V. This voltage is sufficient to provide the 9V  
gate drive to the MOSFET that is required to achieve a low  
RDS(ON). Since the low side synchronous FET is referenced to  
ground (see Figure 4), boosting the gate drive voltage is not  
needed and the VCCP power pin can be tied to VCC.  
Refer to Typical Operating Characteristics of the RC5040  
data sheet for a full load VCCQP waveform.  
The RC5040 and RC5042 Power Good function has been  
designed according to Intel’s Pentium Pro DC-DC converter  
specification. The Power Good function provides a constant  
voltage monitor on the VFB pin. The internal circuitry of the  
converter compares the VFB signal to the VREF voltage and  
outputs an active-low interrupt signal to the CPU when the  
power supply voltage exceeds ±7% of its nominal setpoint.  
The Power Good flag provides no other control function to  
the RC5040.  
Output Enable (OUTEN)  
Intel specifications state that the DC-DC converter should  
accept an open collector signal for controlling the output  
voltage. A logic LOW for this signal disables the output volt-  
age. When disabled, the PWRGD output is in the low state.  
This feature is available for the RC5040 only.  
Upgrade Present (UP#)  
Intel specifications state that the DC-DC converter must  
accept an open collector signal that indicates the presence of  
an upgrade processor. The typical state is high (for a stan-  
dard P6 processor). When the signal is low or in theground  
state (for the OverDrive processor), the output voltage must  
be disabled unless the converter can supply the OverDrive  
processor’s power requirements. When disabled, the  
PWRGD output must be in the low state. Because the  
RC5040 and RC5042 can supply the OverDrive processor  
requirements, the UP# signal is not required.  
Internal Voltage Reference  
The reference used in the RC5040 is a precision band-gap  
voltage reference, with internal resistors precisely trimmed  
to provide a near zero temperature coefficient, TC. Added to  
the reference voltage is the output from a 4-bit DAC. The  
DAC is provided meet Pentium Pro specifications, requiring  
a programmable converter output via a 4-bit voltage identifi-  
cation (VID) code. This code scales the output voltage from  
2.0V (no CPU) to 3.5V in 100mV increments. To guarantee  
stable operation under all loads, a 10Kpull-up resistor and  
0.1µF of decoupling capacitance should be connected to the  
VREF pin. No load should be imposed on this pin.  
4
APPLICATION NOTE  
AN42  
In general, a lower operating frequency increases the peak  
Over-Voltage Protection  
ripple current flowing through the output inductor, allowing  
the use of a larger inductor value. Operation at lower fre-  
quencies increases the amount of energy storage that the  
bulk output capacitors must provide during load transients  
that occur due to the slower loop response of the controller.  
The RC5040 and RC5042 constantly monitor the output  
voltage for protection against over voltage. If the voltage at  
the VFB pin exceeds 20% of the selected program voltage,  
an over-voltage condition is assumed, and the controller dis-  
ables the output drive signal to the external MOSFET(s).  
In addition, note that the efficiency losses due to switching  
are relatively fixed per switching cycle. Therefore, as the  
switching frequency increases, the contribution toward effi-  
ciency due to switching losses also increases.  
Short Circuit Protection  
A current sense methodology is implemented to disable the  
output drive signal to the MOSFET(s) when an over-current  
condition is detected. The voltage drop created by the output  
current flowing across a sense resistor is presented to an  
internal comparator. When the voltage developed across the  
sense resistor exceeds the comparator threshold voltage,  
the controller disables the output drive signal to the  
MOSFET(s).  
RC5040 has an optimal operating frequency of 650KHz.  
This frequency allows the use of smaller inductive and  
capacitive components while optimizing peak efficiency  
under all operating conditions.  
Design Considerations and  
Component Selection  
The DC-DC converter returns to normal operation after the  
fault has been removed, for either an over voltage or a short  
circuit condition.  
Application Circuits  
Oscillator  
Figure 3 illustrates a typical non-synchronous application  
using the RC5040. Figure 4 shows a typical synchronous  
application using the RC5040, and Figure 5 shows a typical  
non-synchronous application using the RC5042.  
The RC5040 oscillator section is implemented using a  
fixed current capacitor charging configuration. An external  
capacitor (CEXT) is used to preset the oscillator frequency  
between 200KHz and 1MHz. This allows maximum flexibil-  
ity in setting the switching frequency and in choosing exter-  
nal components.  
L2  
VCC  
2.6µH  
C5  
C4  
C3  
C1  
C2  
0.1µF  
0.1µF  
1000µF 1000µF  
1000µF  
DS2  
C9  
C8  
1N5817  
0.1µF  
0.1µF  
M1  
M2  
C12  
2SK1388  
11  
12  
10  
9
8
7
6
5
4
3
2
1
1µF  
2SK1388  
L1  
R
13  
14  
15  
SENSE  
R7  
VO  
10K  
1.3µH  
8mΩ  
C6  
RC5040  
4.7µF  
16  
17  
18  
19  
20  
VREF  
GND  
C7  
DS1  
MBR1545CT  
0.1µF  
C
EXT  
39pF  
10K  
10K  
10K  
10K  
R1  
VID3  
VID2  
VCC  
R2  
R3  
R4  
R6  
10K  
PWRGD  
VID1  
VID0  
65-AP42-03  
C11  
0.22µF  
VCC  
R5  
10K  
OUTEN  
C10  
0.1µF  
Figure 3. Non-Synchronous DC-DC Converter Application Schematic Using RC5040  
5
AN42  
APPLICATION NOTE  
L2  
VCC  
2.6µH  
C4  
C1  
C2  
C5  
C3  
0.1µF  
0.1µF  
1000µF  
1000µF  
1000µF  
DS2  
C9  
C8  
1N5817  
0.1µF  
0.1µF  
M1  
M2  
C12  
2SK1388  
11  
12  
13  
14  
15  
10  
9
8
7
6
5
1µF  
2SK1388  
L1  
R
SENSE  
R7  
VO  
10K  
1.3µH  
8mΩ  
C6  
4.7µF  
RC5040  
16  
17  
18  
19  
20  
VREF  
4
3
2
1
M3  
2SK1388  
C7  
DS1  
1N5817  
0.1µF  
GND  
C
EXT  
39pF  
10K  
10K  
10K  
10K  
R1  
VID3  
VCC  
R2  
R3  
R4  
R6  
VID2  
10K  
PWRGD  
VID1  
VID0  
65-AP42-04  
C11  
VCC  
0.22µF  
R5  
10K  
OUTEN  
C10  
0.1µF  
Figure 4. Synchronous DC-DC Converter Application Schematic Using RC5040  
L2  
2.6µH  
VCC  
C4  
C5  
C3  
C1  
C2  
0.1µF  
0.1µF  
1000µF 1000µF  
1000µF  
DS2  
C9  
C8  
1N5817  
0.1µF  
0.1µF  
M2  
C12  
2SK1388  
M1  
1µF  
2SK1388  
L1  
R
9
10  
11  
12  
8
7
6
5
SENSE  
R7  
VO  
1.3µH  
10K  
8mΩ  
C6  
4.7µF  
VREF  
GND  
RC5042  
4
3
2
1
13  
14  
15  
16  
DS1  
C7  
MBR1545CT  
0.1µF  
C
EXT  
39pF  
10K  
10K  
10K  
10K  
R1  
R2  
R3  
R4  
VID3  
VCC  
R6  
10K  
VID2  
65-AP42-05  
PWRGD  
VID1  
VID0  
C11  
0.22µF  
VCC  
C10  
0.1µF  
Figure 5. Non-Synchronous DC-DC Converter Application Schematic Using RC5042  
6
APPLICATION NOTE  
AN42  
• Power package with low thermal resistance  
• Drain current rating of 20A minimum  
• Drain-Source voltage > 15V.  
MOSFET Selection  
This application requires the use of N-channel, Logic Level  
Enhancement Mode Field Effect Transistors. The desired  
characteristics of these components are:  
The on-resistance (R ) is the main parameter for MOS-  
DS,ON  
FET selection. It determines the MOSFET’s power dissipa-  
tion, thus significantly affecting the efficiency of the  
converter. Several suitable MOSFETs are shown in Table 5.  
• Low Static Drain-Source On-Resistance  
R
< 37 m(lower is better)  
DS,ON  
• Low gate drive voltage, V 4.5V  
GS  
Table 5. MOSFET Selection Table  
R
DS,ON  
(m)  
Thermal  
1
Manufacturer & Model #  
Conditions  
T = 25°C  
Typ.  
25  
Max.  
P ackage Resistance  
Fuji  
2SK1388  
V
D
= 4V  
37  
20  
34  
15  
TO-220  
Φ
Φ
= 75  
= 50  
GS  
I = 17.5A  
J
JA  
JA  
T = 125°C  
J
37  
Siliconix  
SI4410DY  
V
= 4.5V  
T = 25°C  
J
16.5  
28  
SO-8  
(SMD)  
GS  
I = 5A  
D
T = 125°C  
J
National Semiconductor  
NDP706AL  
V
= 5V  
T = 25°C  
J
13  
TO-220  
Φ
= 62.5  
JA  
GS  
I = 40A  
Φ
= 1.5  
D
JC  
NDP706AEL  
National Semiconductor  
NDP603AL  
T = 125°C  
20  
31  
42  
22  
33  
6
24  
40  
54  
25  
40  
9
J
V
D
= 4.5V  
T = 25°C  
J
TO-220  
TO-220  
TO-263  
Φ
= 62.5  
JA  
GS  
I = 10A  
T = 125°C  
J
Φ
= 2.5  
JC  
National Semiconductor  
NDP606AL  
V
= 5V  
T = 25°C  
J
Φ
JA  
= 62.5  
GS  
I = 24A  
D
T = 125°C  
J
Φ
= 1.5  
JC  
Motorola  
V
= 5V  
T = 25°C  
J
Φ
JA  
= 62.5  
GS  
I = 37.5A  
D
2
MTB75N03HDL  
Int. Rectifier  
T = 125°C  
J
9.3  
14  
28  
46  
19  
31  
(D PAK)  
Φ
= 1.0  
JC  
V
= 5V  
T = 25°C  
J
TO-220  
TO-220  
Φ
JA  
= 62.5  
GS  
I = 31A  
D
IRLZ44  
T = 125°C  
J
Φ
= 1.0  
JC  
Int. Rectifier  
V
= 4.5V  
T = 25°C  
J
Φ
JA  
= 62.5  
GS  
I = 28A  
D
IRL3103S  
T = 125°C  
J
Φ
= 1.0  
JC  
Note:  
1. R  
) values at Tj = 125°C for most devices were extrapolated from the typical operating curves supplied by the manufac-  
DS(ON  
turers and are approximations only.  
Two MOSFETs in Parallel  
We recommend two MOSFETs used in parallel instead of a  
single MOSFET. The following significant advantages are  
realized using two MOSFETs in parallel:  
• No added heat sink required.  
With the power dissipation down to around one watt and  
with MOSFETs mounted flat on the motherboard, no  
external heat sink is required. The junction-to-case  
thermal resistance for the MOSFET package (TO-220) is  
typically at 2°C/W and the motherboard serves as an  
excellent heat sink.  
Significant reduction of power dissipation.  
Maximum current of 14A with one MOSFET:  
2
P
= (I R  
)(Duty Cycle) =  
DS,ON  
MOSFET  
2
• Higher current capability.  
(14) (0.050*)(3.3+0.4)/(5+0.4-0.35) = 7.2 W  
With thermal management under control, this on-board  
DC-DC converter can deliver load currents up to 14.5A  
with no performance or reliability concerns.  
With two MOSFETs in parallel:  
2
P
= (I R  
)(Duty Cycle) =  
DS,ON  
MOSFET  
2
(14/2) (0.037*)(3.3+0.4)/(5+0.4-0.35) = 1.3W/FET  
*
Note: R  
increases with temperature. Assume R  
= 25mΩ  
can easily increase to 50mat high temperature  
DS,ON  
DS,ON  
at 25°C. R  
DS,ON  
when using a single MOSFET. When using two MOSFETs in  
parallel, the temperature effects should not cause the R  
to rise  
DS,ON  
above the listed maximum value of 37m.  
7
AN42  
APPLICATION NOTE  
12V Gate Bias  
MOSFET Gate Bias  
Figure 7 illustrates how an external 12V source can be used  
to bias VCCQP. A 47 resistor is used to limit the transient  
current into the VCCQP pin, and a 1µF capacitor filter is  
used to filter the VCCQP supply. This method provides a  
The MOSFET(s) can be biased using one of two methods:  
Charge Pump or 12V Gate Bias.  
Charge Pump (or Bootstrap)  
higher gate bias voltage (V ) to the MOSFET, and there-  
GS  
Figure 6 employs a charge pump to provide the MOSFET  
gate bias. The charge pump capacitor, CP, is used as a flying  
capacitor to boost the voltage of the RC5040 or RC5042 out-  
put driver. When the MOSFET switches off, the source of the  
MOSFET is at -0.6V. VCCQP is charged through the Schot-  
tky diode to 4.5V. Thus, the capacitor CP is charged to 5V.  
When the MOSFET turns on, the source of the MOSFET is  
at approximately 5V. The capacitor voltage follows, and  
hence provides a voltage at VCCQP equal to 10V. The Schot-  
tky is required to provide the charge path when the MOSFET  
is off, and then reverses bias when the VCCQP goes to 10V.  
The capacitor CP needs to be a high Q and high frequency  
capacitor. A 1µF ceramic capacitor is recommended here.  
fore reduces the R  
DS,ON  
MOSFET. Figure 8 illustrates how R  
and resulting power loss within the  
decreases dra-  
DS,ON  
matically as V increases. A 6.2V Zener (DS2) is used to  
GS  
clamp the voltage at V  
CCQP  
to a maximum of 12V and  
ensure that the absolute maximum voltage of the IC is not  
exceeded.  
Warning: The 12V Gate Bias method applies only to the  
RC5042. The RC5040 has not been designed to accept an  
external 12V gate bias voltage, and may be damaged if  
this method is used.  
+5V  
+5V  
47  
+12V  
D1  
DS2  
6.2V  
VCCQP  
M1  
VCCQP  
M1  
HIDRV  
HIDRV  
CP  
L1  
RS  
L1  
RS  
VO  
PWM/PFM  
Control  
VO  
PWM/PFM  
Control  
CB  
DS1  
CB  
DS1  
65-AP42-06  
65-AP42-07  
Figure 6. Charge Pump Configuration  
Figure 7. 12V Gate Bias Configuration  
0.1  
0.09  
0.08  
0.07  
0.06  
0.05  
0.04  
0.03  
0.02  
0.01  
0
R(DS)Fuji  
R(DS)Fuji  
R(DS)706A  
R(DS)-706AEL  
1.5  
2
2.5  
3
3.5  
4
5
6
7
8
9
10  
11  
Gate-Source Voltage, V  
(V)  
GS  
Figure 8. R  
DS,ON  
vs. V for Selected MOSFETs  
GS  
8
APPLICATION NOTE  
AN42  
Converter Efficiency  
Losses due to parasitic resistance in the switches, inductor,  
and sense resistor dominate at high load-current levels. The  
major loss mechanisms under heavy loads, in order of  
importance, are:  
• Sense Resistor Losses  
• Gate-Charge Losses  
• Diode-Conduction Losses  
• Transition Losses  
• Input Capacitor Losses  
• Losses Due to the Operating Supply Current of the IC.  
2
• MOSFET I R Losses  
• Inductor Losses  
Efficiency of the converter under heavy loads can be calculated as follows:  
× V  
P
I
OUT  
OUT  
OUT  
Efficiency = ------------- = -------------------------------------------------------- ,  
p
I
× V  
+ P  
OUT LOSS  
IN  
OUT  
where P  
= PD  
+ PD  
+ PD  
+ PD  
+ PD  
+ PD  
+ PD  
+ PD  
CAP IC  
LOSS  
MOSFET  
INDUCTOR  
RSENSE  
GATE  
DIODE  
TRAN  
Design Equations:  
V
+ V  
D
2
OUT  
(1) PD  
(2) PD  
= I  
× R  
× DutyCycle , whereDutyCycle = -----------------------------------------  
DS,ON  
MOSFET  
OUT  
V
+ V – V  
SW  
IN  
D
2
= I  
× R  
INDUCTOR  
INDUCTOR  
OUT  
2
(3) PD  
(4) PD  
= I  
× R  
OUT SENSE  
RSENSE  
= q  
× f × 5V , where q  
is the gate charge and f is the switching frequency  
GATE  
GATE  
GATE  
(5) PD  
= V × I (1 – DutyCycle)  
DIODE  
f
D
2
IN  
V
× C  
× I  
× f  
LOAD  
RSS  
(6) PD  
(7) PD  
= ------------------------------------------------------------- , where C  
is the reverse transfer capacitance of the high-side MOSFET.  
RSS  
TRAN  
CAP  
I
DRIVE  
2
= I  
× ESR  
RMS  
(8) PD = V × I  
IC  
CC  
CC  
Example:  
3.3 + 0.5  
DutyCycle = ----------------------------- = 0.73  
5 + 0.5 – 0.3  
2
PD  
PD  
= 10 × 0.030 × 0.73 = 2.19W  
MOSFET  
2
= 10 × 0.010 = 1W  
INDUCTOR  
2
PD  
PD  
PD  
= 10 × 0.0065 = 0.65W  
RSENSE  
= CV × f × 5V = 1.75nf × (9 – 1)V × 650Khz × 5V = 0.045W  
GATE  
= 0.5 × 10(1 – 0.73) = 1.35W  
DIODE  
2
5 × 400pf × 10 × 650khz  
----------------------------------------------------------------  
0.7A  
PD  
PD  
=
0.010W  
TRAN  
2
= (7.5 – 2.5) × 0.015 = 0.37W  
CAP  
PD = 0.2W  
IC  
9
AN42  
APPLICATION NOTE  
PD  
= 2.19W + 1.0W + 0.65W + 0.045W + 1.35W + 0.010W + 0.37W + 0.2W = 5.815W  
LOSS  
3.3 × 10  
---------------------------------------  
Efficiency =  
85%  
3.3 × 10 + 5.815  
Table 6. RC5040 and RC5042 Short Circuit Comparator  
Threshold Voltage  
Selecting the Inductor  
Selecting the right inductor component is critical in the  
DC-DC converter application. The inductor’s critical param-  
eters to consider are inductance (L), maximum DC current  
Short Circuit Comparator  
V
(mV)  
threshold  
(I ), and coil resistance (R ).  
O
l
Typical  
Minimum  
Maximum  
120  
100  
The inductor core material is crucial in determining the  
amount of current it can withstand. As with all engineering  
designs, tradeoffs exist between various types of core mate-  
rials. In general, Ferrites are popular due to their low cost,  
low EMI properties, and high frequency (>500KHz) charac-  
teristics. Molypermalloy powder (MPP) materials exhibit  
good saturation characteristics, low EMI, and low hysteresis  
losses; however, they tend to be expensive and more effec-  
tively utilized at operating frequencies below 400KHz.  
140  
When designing the external current sense circuitry, pay  
careful attention to the output limitations during normal  
operation and during a fault condition. If the short circuit  
protection threshold current is set too low, the converter may  
not be able to continuously deliver the maximum CPU load  
current. If the threshold level is too high, the output driver  
may not be disabled at a safe limit and the resulting power  
dissipation within the MOSFET(s) may rise to destructive  
levels.  
Another critical parameter is the DC winding resistance of  
the inductor. This value should typically be as low as possi-  
ble because the power loss in DC resistance degrades the  
The design equation used to set the short circuit threshold  
limit is as follows:  
2
efficiency of the converter by P  
LOSS  
= I x R . The value  
O
l
of the inductor is a function of the oscillator duty cycle  
(T ) and the maximum inductor current (I ). I can be  
V
th  
ON  
PK PK  
--------  
R
=
, where: I = output short circuit current  
SENSE  
SC  
calculated from the relationship:  
I
SC  
(I – I  
)
min  
pk  
V
– V  
– V  
SW D  
I
I  
= I  
+ ----------------------------  
Load, max  
IN  
SC  
inductor  
-----------------------------------------  
I
= I  
+
T
ON  
2
PK  
MIN  
L
where I and I  
are peak ripple currents and  
is the maximum output load current.  
pk min  
Where T  
ON  
forward voltage of diode DS1.  
is the maximum duty cycle and V is the  
D
I
load, max  
You must also take into account the current (I –I ), or  
pk min  
The inductor value can be calculated using the following  
relationship:  
the ripple current flowing through the inductor under normal  
operation. Figure 9 illustrates the inductor current waveform  
for the RC5040 and RC5042 DC-DC converters at maxi-  
mum load.  
V
– V  
– V  
SW O  
IN  
-----------------------------------------  
L =  
T
ON  
I
– I  
MIN  
PK  
I
pk  
Where V  
(R x I ) is the drain-to-source voltage of  
SW DS,ON  
O
M1 when it is turned on.  
I
(I – I  
pk  
)/2  
min  
Implementing Short Circuit Protection  
ILOAD, MAX  
t
I
min  
Intel currently requires all power supply manufacturers  
to provide continuous protection against short circuit  
conditions that may damage the CPU. To address this  
requirement, Raytheon Electronics has implemented a cur-  
rent sense methodology on the RC5040 and RC5042 con-  
trollers. This methodology limits the power delivered to the  
load during an overcurrent condition. The voltage drop cre-  
ated by the output current flowing across a sense resistor is  
presented to one terminal of an internal comparator with  
hysterisis. The other comparator terminal has a threshold  
voltage, nominally 120mV. Table 6 states the limits for the  
comparator threshold of the switching regulator:  
TON  
TOFF  
T = 1/f  
s
Figure 9. Typical DC-DC Converter  
Inductor Current Waveform  
The calculation of this ripple current is as follows:  
(V –V – V (V + V )  
)
(I – I  
)
min  
IN  
SW  
OUT  
OUT  
D
pk  
------------------------------------------------ ---------------------------------------------  
T
---------------------------- =  
×
L
(V – V + V )  
IN SW  
2
D
10  
APPLICATION NOTE  
AN42  
(IPK – Imin  
)
where:  
ISC Iinductor = ILoad, max + ----------------------------- = 14.5 + 1 = 15.5A  
2
V
V
V
= Input Voltage to the Converter  
= Voltage Across the MOSFET = I  
LOAD  
= Forward Voltage of the Schottky Diode  
IN  
SW  
D
For continuous operation at 14.5A, the short circuit detection  
threshold must be at least 15.5A.  
x R  
DS,ON  
T = The Switching Period of the Converter = 1/fS,  
Where f = Switching Frequency.  
The next step is to determine the value of the sense resistor.  
Including tolerance, the sense resistor value can be approxi-  
mated as follows:  
S
For an input voltage of 5V, an output voltage of 3.3V, an  
Vth,min  
Vth,min  
inductor value of 1.3µH, and a switching frequency of  
----------------  
ISC  
----------------------------------  
1.0 + ILoad,max  
RSENSE  
=
× (1 – TF) =  
× (1 – TF)  
650KHz (using C  
EXT  
= 39pF), the inductor current can be  
calculated as follows:  
where TF = Tolerance Factor for the sense resistor.  
(I – I  
)
min  
(5.0 – 14.5 × 0.037 – 3.3)  
pk  
-------------------------------------------------------------  
---------------------------- =  
×
–6  
2
Several different types of sense resistors exist. Table 7  
describes tolerance, size, power capability, temperature  
coefficient and cost of various sense resistors.  
1.3 × 10  
(3.3 + 0.5)  
1
-------------------------------------------------------------- -----------------------  
×
= 1.048A  
3
(5.0 – 14.5 × 0.037 + 0.5)  
650 × 10  
Therefore, for a continued load current of 14.5A, the peak  
current through the inductor, I , is found to be:  
pk  
Table 7. Comparison of Sense Resistors  
Discrete Metal  
Strip Surface  
Mount Resistor  
(Dale)  
Discrete  
CuNi Alloy  
Wire Resistor  
(Copel)  
Discrete Iron  
Alloy  
Resistor (IRC)  
Discrete MnCu  
Alloy Wire  
Resistor  
Motherboard  
Trace Resistor  
Description  
Tolerance  
Factor (TF)  
±29%  
±5%  
(±1% available)  
±1%  
±10%  
±10%  
Size  
(L x W x H)  
2" x 0.2" x 0.001" 0.45" x 0.065" x 0.25" x 0.125" x 0.200" x 0.04" x  
0.200" x 0.04" x  
0.100"  
(1 oz Cu trace)  
0.200"  
0.025"  
0.160"  
Power capability  
> 50A/in  
1 watt  
1 watt  
1 watt  
1 watt  
(3W and 5W  
available)  
Temperature  
Coefficient  
+4,000 ppm  
+30 ppm  
±75 ppm  
±30 ppm  
±20 ppm  
Cost  
Low  
$0.31  
$0.47  
$0.09  
$0.09  
@10,000 piece  
included in  
motherboard  
Refer to Appendix A for Directory of component suppliers  
Based on the Tolerance in the above table, for embedded PC  
V
th,min  
----------------------------------------  
× (1 – TF) =  
R
=
trace resistor and for I  
= 14.5A:  
load,max  
SENSE  
1.0A + I  
Load, max  
V
th,min  
----------------------------------------  
R
=
× (1 – TF) =  
SENSE  
1.0A + I  
Load, max  
100mV  
---------------------------------  
× (1 – 5%) = 6.1mΩ  
1.0A + 14.5A  
100mV  
---------------------------------  
1.0A + 14.5A  
× (1 – 29%) = 4.6mΩ  
For user convenience, Table 8 lists the recommended values  
for sense resistor at various load currents using an embedded  
PC trace resistor or discrete resistor.  
For a discrete resistor and I  
= 14.5A:  
load, max  
11  
AN42  
APPLICATION NOTE  
Table 8. R  
for various load currents  
Table 9 is a summary of tolerances for the Embedded PC  
Trace Resistor.  
sense  
R
R
SENSE  
SENSE  
I
PC Trace  
Resistor (m)  
Discrete  
Resistor (m)  
Load,max  
Table 9. Summary PC Trace Resistor Tolerance  
(A)  
Tolerance due to sheet resistivity variation  
Tolerance due to L/W error  
16%  
1%  
10.0  
11.2  
12.4  
13.9  
14.0  
14.5  
6.5  
5.8  
5.3  
4.8  
4.7  
4.6  
8.6  
7.8  
7.1  
6.4  
6.3  
6.1  
Tolerance due to temperature variation  
Total Tolerance for PC Trace Resistor  
12%  
29%  
Design rules for using an embedded resistor  
The basic equation for laying an embedded resistor is:  
L
L
W × t  
W
-------------  
R = ρ ×  
t
Discrete Sense Resistor  
Discrete iron alloy resistors come in a variety of tolerances  
and power ratings, and are ideal for precision implementa-  
tions. Either an MnCu alloy wire resistor or an CuNi alloy  
wire resistor is ideal for a low cost implementation.  
where ρ is the Resistivity (W-mil), L is the Length (mils), W  
is the Width (mils), and t is the Thickness (mils).  
For 1oz copper, t = 1.35 mils, ρ = 717.86 µΩ-mil,  
1 L/1 W = 1 Square ( ).  
For example, you can layout a 5.30membedded sense  
Embedded Sense Resistor (PC Trace Resistor)  
resistor. From Equations above,  
Embedded PC trace resistors have the advantage of almost  
zero cost implementation. However, the value of the PC  
trace resistors have large variations. Embedded resistors  
have 3 major error sources: the sheet resistivity of the inner  
layer, the mismatch due to L/W, and the temperature varia-  
tion of the resistor. When laying out embedded sense resis-  
tors, consider all error sources described as follows:  
I
10  
0.05  
L
W = --------- = --------- = 200mil  
0.05  
R × W × t  
0.00530 × 200 × 1.35  
L = ----------------------- = --------------------------------------------------- = 2000mi  
ρ
717.86  
L/W = 10 .  
Therefore, to model 5.30menbedded resistor, you need  
Sheet resistivity.  
W = 200 mils, and L = 2000 mils. See Figure 10.  
For 1 ounce copper, the thickness variation is typically  
between 1.15 mil and 1.35 mil. Therefore, the error due to  
sheet resistivity is (1.35 – 1.15)/1.25 = 16%.  
1
1
1
1
1
1
1
1
1
1
W = 200 mils  
Mismatch due to L/W.  
The error in L/W is dictated by the geometry and the  
power dissipation capability of the sense resistor. The  
sense resistor must be able to handle the load current and,  
therefore, requires a minimum width, calculated as  
follows:  
L = 2000  
Figure 10. 5.30mSense Resistor (10 )  
You can also implement the sense resistor in the following  
manner. Each corner square is counted as 0.6 square since  
the current flowing through the corner square does not flow  
uniformly, concentrated towards the inside edge. This is  
shown in Figure 11.  
I
L
W = ---------  
0.05  
where W is the minimum width required for proper power  
dissipation (mils), and I is the load current in Amps.  
L
1
1
1
1
1
1
.6  
.6  
1
For a load current of 15A, the minimum width required is  
300mils, which reflects a 1% L/W error.  
1
.8  
Thermal Considerations.  
2
The I R power losses cause the surface temperature of the  
Figure 11. 5.30mSense Resistor (10 )  
resistor to increase along with its resistance value. In  
addition, ambient temperature variations add the change  
in resistor value:  
A Resign Example Combining an Embedded Resistor  
with a Discrete Resistor  
R = R [1 + α (T – 20)]  
20  
20  
For low cost implementation, the embedded PC trace resistor  
is the most desirable alternative, but, as discussed earlier, the  
wide tolerance (±29%) presents a challenge. In addition,  
changing CPU requirements may force the maximum load  
where R is the resistance at 20°C, α = 0.00393/ °C,T  
20  
20  
is the operating temperature, andR is the desired value.  
For temperature T = 50°C, the %R change = 12%.  
12  
APPLICATION NOTE  
AN42  
Embedded Sense Resistor  
IFBH  
MnCu Discrete  
Resistor  
R21  
R22  
IFBL  
Output Power  
Plane (Vout)  
R-r  
R
R+r  
Figure 12. Short Circuit Sense Resistor Design Using PC Trace Resistor and Optional Discrete Sense Resistor  
currents to change. Therefore, combining an embedded  
resistor with a discrete resistor may be a desirable option.  
This section discusses a design that provides flexibility and  
addresses wide tolerances. Refer to Figure 12.  
The converter exhibits at normal load regulation until the  
voltage across the resistor reaches the internal short circuit  
threshold of 120mV. At this point, the internal  
comparator trips and signals the controller to turn off the  
gate drive to the power MOSFET. This causes a drastic  
reduction in the output voltage as the load regulation col-  
lapses into the short circuit control mode. The output voltage  
does not return to its nominal value until the output short cir-  
cuit current is reduced to within the safe range for the DC-  
DC converter.  
In this design, the user has the option to choose either an  
embedded or a discrete MnCu sense resistor. To use the dis-  
crete sense resistor, populate R21 with a shorting bar (zero  
Ohm resistor) for a proper Kelvin connection and add the  
MnCu sense resistor. To use the embedded sense resistor,  
populate R22 with a shorting bar for a Kelvin connection.  
The embedded sense resistor allows you to choose a plus or a  
minus delta resistance tap to offset any large sheet resistivity  
change.  
Power Dissipation Consideration During a  
Short Circuit Condition  
The RC5040 and RC5042 controllers respond to an output  
short circuit by drastically changing the duty cycle of the  
gate drive signal to the power MOSFET. In doing this, the  
power MOSFET is protected from over-stress and eventual  
destruction. Figure 14A shows the gate drive signal of a typ-  
ical RC5040 operating in continuous mode with a load cur-  
rent of 10A. The duty cycle is then set by the ratio of the  
input voltage to the output voltage. If the input voltage is 5V  
and the output voltage is 3.1V, the ratio of Vout/ Vin is 62%.  
Figure 14B shows the result of the RC5040 going into its  
short circuit mode when the duty cycle is around 20%. Cal-  
culating the power on the MOSFET at each condition on the  
graph in Figure 13 shows how the protection scheme works.  
The power dissipated in the MOSFET at normal operation  
for a load current of 14.5A, is given by:  
In this design, the center tap yields 6m, and the left or the  
right tap yield 6.7 or 5.3 mΩ, respectively.  
RC5040 and RC5042 Short Circuit Current  
Characteristics  
The RC5040 and RC5042 have a short circuit current char-  
acteristic that includes a hysteresis function. This function  
prevents the DC-DC converter from oscillating in the event  
of a short circuit. Figure 13 shows the typical characteristic  
of the DC-DC converter using a 6.5 msense resistor.  
3.5  
˙
2 × .037 × .62 = 1.2W  
3.0  
2.5  
2.0  
1.5  
1.0  
14.5  
2
PD = I2 × RON × DutyCycle =  
---------  
for each MOSFET.  
The power dissipated in the MOSFET at short circuit  
condition for a peak short current of 20A, is given by:  
0.5  
0
2
20  
2
-----  
P
=
× .037 × .2 = 0.74W  
D
0
5
10  
15  
20  
25  
for each MOSFET.  
Output Current  
Figure 13. RC5040/RC5042 Short Circuit Characteristic  
Thus, the MOSFET is not being over-stressed during a short  
circuit condition.  
13  
AN42  
APPLICATION NOTE  
P
= I  
× V × (1 – DutyCycle) =  
F, ave  
D, Diode  
F
14 × 0.45 × 0.8 5W  
Thus for the Schottky diode, the thermal dissipation during  
a short circuit is greatly magnified and requires that the  
thermal dissipation of the diode be properly managed by the  
appropriate choice of a heat sink. In order to protect the  
Schottky from being destroyed in the event of a short, we  
should limit the junction temperature to less than 130°C.  
Using the equation for maximum junction temperature,  
we can arrive at the thermal resistance required below:  
T
– T  
A
J(max)  
R
P
= -------------------------------  
D
ΘJA  
Figure 14A. V  
CCQP  
Output Waveform for Normal  
= 3.3V@10A  
Operation Condition with V  
Assuming that the ambient temperature is 50°C, we get:  
– T  
out  
T
130 – 50  
J(max)  
A
R
= ------------------------------- = -------------------- = 16°C W  
ΘJA  
P
5
D
Thus we need to provide for a heat sink that will give the  
Schottky diode a thermal resistance of at least 16°C/W or  
lower in order to protect the device during an indefinite  
short.  
In summary, with proper heat sink, the Schottky diode is not  
being over stressed during a short circuit condition.  
Schottky Diode Selection  
The application circuits of Figures 3, 4, and 5 show two  
Schottky diodes, DS1 and DS2. In synchronous mode, DS1  
is used in parallel with M3 to prevent the lossy diode in the  
FET from turning on. In non-synchronous mode, DS1 is  
used as a flyback diode to provide a constant current path for  
the inductor when M1 is turned off.  
Figure 14B. V  
CCQP  
Output Waveform for  
Output Shorted to Ground  
The Schottky diode has a power dissipation consideration  
during the short circuit condition. During normal operation,  
the diode dissipates power when the power MOSFET is off.  
The power dissipation is given by:  
The Schottky diode DS2 serves a dual purpose. As config-  
ured in Figures 3, 4, and 5, DS2 allows the VCCQP pin on  
the RC5040 to be bootstrapped up to 9V using capacitor  
C12. When the lower MOSFET M3 is turned on, one side of  
capacitor C12 is connected to ground while the other side of  
the capacitor is being charged up to voltage VIN – VD  
through DS2. The voltage that is then applied to the gate of  
the MOSFET is VCCQP – VSAT, or typically around 9V.  
DS2 also provides correct sequencing of the various supply  
voltages by assuring that VCCQP is not enabled before the  
other supplies.  
P
= I × V × (1 – DutyCycle) =  
D, Diode  
F
F
14.5 × 0.5V × (1 – 0.62) = 2.75W  
In short circuit mode, the duty cycle is dramatically reduced  
to approximately 20%. The forward current during a short  
circuit condition decays exponentially through the inductor.  
The power dissipated on the diode during the short circuit  
condition, is approximated by:  
A vital selection criteria for DS1 and DS2 is that they exhibit  
a very low forward voltage drop, as this parameter can  
directly affect the regulator efficiency. Table 10 lists several  
suitable Schottky diodes. Note that the MBR2015CTL has a  
very low forward voltage drop. This diode is ideal for appli-  
cations where output voltages less than 2.8V are required.  
1
1.5us  
-------------  
1.3us  
-----------  
L R  
I
I
= I × e  
= 20A × e  
7.9A  
F, ending  
sc  
(20A + 7.9A) ⁄ 2 14A  
F, ave  
14  
APPLICATION NOTE  
AN42  
With this in mind, correct calculation of the output capaci-  
Table 10. Schottky Diode Selection Table  
tance is crucial to the performance of the DC-DC converter.  
The output capacitor determines the overall loop stability,  
output voltage ripple, and load transient response. The calcu-  
lation is as follows:  
Manufacturer  
Model #  
Forward Voltage  
Conditions  
VF  
Philips  
PBYR1035  
IF = 20A; Tj=25°C  
IF = 20A; Tj=125°C  
< 0.84v  
< 0.72v  
I × T  
O
Motorola  
MBR2035CT IF = 20A; Tj=125°C  
IF = 20A; Tj=25°C  
< 0.84v  
< 0.72v  
C(µF) = -------------------------------------  
V – I × ESR  
O
Motorola  
MBR1545CT IF = 15A; Tj=125°C  
IF = 15A; Tj=25°C  
< 0.84v  
< 0.72v  
where V is the maximum voltage deviation due to load  
transients, T is the reaction time of the power source, and  
I
O
is the output load current. V is the loop response time of  
Motorola  
MBR2015CTL IF = 20A; Tj=150°C  
IF = 20A; Tj=25°C  
< 0.58v  
< 0.48v  
the RC5040 and RC5042, approximately 8µs.  
For I = 10A and V = 165mV, the bulk capacitance  
O
Output Filter Capacitors  
required can be approximated as follows:  
Output ripple performance and transient response are  
functions of the filter capacitors. Since the 5V supply of a PC  
motherboard may be located several inches away from the  
DC-DC converter, the input capacitance can play an impor-  
tant role in the load transient response of the RC5040.  
The higher the input capacitance, the more charge storage is  
available for improving the current transfer through the  
FET(s). Capacitors with low Equivalent Series Resistance  
(ESR) are best for this type of application and can influence  
the converter's efficiency if not chosen carefully. The input  
capacitor should be placed as close to the drain of the FET as  
possible to reduce the effect of ringing caused by long trace  
lengths.  
IO × T  
C(µF) = ------------------------------------- = --------------------------------------------------------- = 1454µF  
V – IO × ESR 165mV – 10A × 11mΩ  
10A × 8µs  
Input filter  
The DC-DC converter design should include an input induc-  
tor between the system +5V supply and the converter input  
as described below. This inductor will serve to isolate the  
+5V supply from noise occurring in the switching portion of  
the DC-DC converter and also to limit the inrush current into  
the input capacitors during power up. An inductor value of  
around 2.5µH is recommended, as illustrated in Figure 15.  
2.5µH  
5V  
Vin  
ESR is the resonant impedance of the capacitor, and it is dif-  
ficult to quantify. Since the capacitor is actually a complex  
impedance device having resistance, inductance, and capaci-  
tance, it is natural for it to have a resonant frequency. As a  
rule, the lower the ESR, the better suited the capacitor is for  
use in switching power supply applications. Many manufac-  
turers do not supply ESR data, but a useful estimate can be  
obtained using the following equation:  
1000µF, 10V  
0.1µF  
Electrolytic  
65-AP42-17  
Figure 15. Input Filter  
DF  
ESR = ------------  
2πfC  
Bill of Materials  
where DF is the dissipation factor of the capacitor, f is the  
operating frequency, and C is the capacitance in farads.  
The Bill of Materials for the application circuits of Figures 2  
through 4 is presented in Table 11.  
Table 11. Bill of Materials for a 14.5A Pentium Pro Motherboard Application  
C4, C5, C7, C8, C9,  
C10  
Panasonic  
ECU-V1H104ZFX  
0.1µF 50V capacitor  
4.7µF 16V capacitor  
39pF capacitor  
C6  
Panasonic  
ECSH1CY475R  
Cext  
Panasonic  
ECU-V1H121JCG  
C12  
C1, C2, C3  
United Chemicon  
LXF16VB102M  
1000µF 6.3V electrolytic  
capacitor 10mm x 20mm  
ESR<0.047Ω  
C11  
Panasonic  
ECU-V1H224ZFX  
0.22µF 50V capacitor  
15  
AN42  
APPLICATION NOTE  
Table 11. Bill of Materials for a 14.5A Pentium Pro Motherboard Application  
C13, C14, C15  
Sanyo  
6MV1500GX  
1500µF 6.3V electrolytic  
ESR < 0.047 Ω  
Vf<0.72V @ If = 15A  
1A, 20V  
capacitor 10mm x 20mm  
DS1  
(note 1)  
Motorola  
MBR1545CT  
Shottky Diode  
DS2  
L1  
General Instruments 1N5817  
Skynet 320-8107  
Schottky Diode  
1.3µH inductor  
2.5µH inductor  
L2*  
Skynet  
320-6110  
*Optional – will help re-  
duce ripple on 5v line  
M1, M2, M3  
(note 2)  
Fuji  
2SK1388  
N-Channel Logic Level  
Enhancement Mode MOSFET  
R
V
< 37m ohm  
DS(ON)  
< 4V, I > 20A  
GS  
D
Rsense  
COPEL  
A.W.G. #18  
6 milliohm CuNi Alloy Wire  
resistor  
R1, R2, R3, R4, R6,  
R7  
Panasonic ERJ-6ENF10.0KV  
10K 5% Resistors  
U1  
Raytheon  
RC5042M or RC5040M  
DC-DC Converter for Pentium  
Pro  
Refer to Appendix A for Directory of component suppliers.  
Notes:  
1. In synchronous mode using the RC5040, a 1A schottky diode (1N5817) may be substituted for the MBR1545CT.  
2. MOSFET M3 is only required for the RC5040 synchronous application.  
PCB Layout Guidelines and Considerations  
PCB Layout Guidelines  
• The CEXT timing capacitor should be surrounded with a  
• Placement of the MOSFETs relative to the RC5040 is  
ground trace. The placement of a ground or power plane  
critical. The MOSFETs (M1 & M2), should be placed  
underneath the capacitor provides further noise isolation,  
such that the trace length of the HIDRV pin to the FET  
and helps to shield the oscillator from the noise on the  
gate is minimized. A long lead length causes high  
PCB. This capacitor should be placed as close to pin 1 as  
amounts of ringing due to the inductance of the trace and  
possible.  
the large gate capacitance of the FET. This noise radiates  
all over the board, and because it is switching at a high  
• Group the MOSFETs, inductor, and Schottky diode as  
voltage and frequency, it is very difficult to suppress.  
close together as possible. This minimizes ringing derived  
from the inductance of the trace and the large gate  
Figure 16 shows an example of proper MOSFET  
capacitance of the FET. Place the input bulk capacitors as  
placement in relation to the RC5040. It also shows an  
close to the drains of MOSFETs as possible. In addition,  
example of problematic placement for the MOSFETs.  
place the 0.1µF decoupling capacitors right on the drain  
of each MOSFET. This helps to suppress some of the high  
In general, noisy switching lines should be kept away  
frequency switching noise on the DC-DC converter input.  
from the quiet analog section of the RC5040. That is,  
traces that connect to pins 12 and 13 (HIDRV and  
• The traces that run from the RC5040 IFB (pin 4) and VFB  
VCCQP) should be kept far away from the traces that  
(pin 5) pins should be run next to each other and be Kelvin  
connect to pins 1 through 5, and pin 16.  
connected to the sense resistor. Running these lines  
together helps to reject some of the common mode noise  
to the RC5040 feedback input. Run the noisy switching  
• Place the 0.1µF decoupling capacitors as close to the  
RC5040 and RC5042 pins as possible. Extra lead length  
signals (HIDRV & VCCQP) on one layer, and use the  
negates their ability to suppress noise.  
inner layers for power and ground only. If the top layer is  
being used to route all of the noisy switching signals, use  
• Each VCC and GND pin should have its own via to the  
the bottom layer to route the analog sensing signals VFB  
appropriate plane on the board to add isolation between  
and IFB.  
pins  
16  
APPLICATION NOTE  
AN42  
Good layout  
Bad layout  
10  
9
10  
9
11 RC5040  
11 RC5040  
12  
13  
12  
13  
8
7
8
7
14  
15  
16  
17  
14  
15  
16  
17  
6
6
5
4
5
4
18  
19  
18  
19  
20  
3
2
3
2
20  
1
1
= “Quiet” Pins  
Figure 16. Example of Proper MOSFETs Placements  
File can be obtained from Raytheon Electronics Semicon-  
ductor Division’s Marketing Department at (415) 966-7819.  
PC Motherboard Layout and Gerber File  
A reference design for motherboard implementation of the  
RC5040 and RC5042 along with the Layout Gerber File and  
Silk Screen are presented below. The actual PCAD Gerber  
17  
AN42  
APPLICATION NOTE  
18  
APPLICATION NOTE  
AN42  
5. Apply load at 1A increments; an active load (HP6060B  
or equivalent) is suggested.  
Guidelines for Debugging and  
Performance Evaluations  
6. In case of poor regulation, refer to the procedures in the  
DebuggingYour First Design Implementation  
Troubleshooting section.  
Use the following procedure to help you debug your design  
implementation:  
Troubleshooting  
1. If no voltage is registered at the output and the circuit is  
not drawing current, look for openings in the connec-  
tions. Check the circuitry versus the schematic, and the  
power supply pins at the device to ascertain that volt-  
age(s) had been applied.  
1. Note the VID pins settings. They tell you what voltage is  
to be expected.  
2. Do not connect any load to the circuit. While monitoring  
the output voltage, apply power to the part with current  
limiting at the power supply. Do this to make sure that  
no catastrophic shorts occur.  
2. If no voltage is registered at the output and the circuit is  
drawing excessive current (>100mA) with no load,  
check for possible shorts. Trace the path of the excessive  
current to determine if the controller is at fault or if the  
excessive current is due to peripheral components.  
3. Ιf proper voltage is not achieved, follow the procedures  
in the Troubleshooting section.  
3. If the output voltage comes near to, but is not, what is  
expected, check the VID inputs at the device pins. The  
part is factory set to correspond to the VID inputs.  
4. After there is proper voltage, increase the current limit-  
ing of the power supply to 16A.  
19  
AN42  
APPLICATION NOTE  
Load Regulation  
4. Premature shut down can be caused by an inappropriate  
value of sense resistor. See the Sense Resistor section.  
VID  
Iload (A)  
0.5  
Vout (V)  
3.0904  
3.0825  
3.0786  
3.0730  
3.0695  
3.0693  
3.0695  
3.0695  
3.0694  
3.0694  
3.0691  
0.70%  
5. A poor load regulation can have many causes. You  
should first check the voltages and signals at the critical  
pins.  
0100  
1.0  
2.0  
6. The VREF pin should be at the voltage set by the VID  
pins. If the power supply pins are correct and the VID  
pins are correct, the VREF should be at the correct volt-  
age.  
3.0  
4.0  
5.0  
7. Next check the oscillator pin. A saw tooth wave at the  
frequency set by the external capacitor should be seen.  
6.0  
7.0  
8. When the VREF and CEXT pins are determined to be  
correct and the output voltage is still incorrect look at  
the waveform at VCCQP. This pin should be swinging  
from ground to +12V (in the +12V application) and  
from slightly below +5V to about +10V (charge pump  
application). If the VCCQP pin is noisy, with ripples and  
overshoots, then the noise may cause the converter to  
function improperly.  
8.0  
9.0  
9.9  
Load Regulation 0.5A – 9.9A  
VID  
Iload (A)  
0.5  
Vout (V)  
3.2805  
3.2741  
3.2701  
3.2642  
3.2595  
3.2597  
3.2606  
3.2611  
3.2613  
3.2611  
3.2607  
3.2599  
3.2596  
3.2596  
0.64%  
0010  
9. Next, look at the HIDRV pin. This pin directly drives the  
gate of the FET. It should provide a gate drive (Vgs) of  
about 5V when turning the FET on. A careful study of  
the layout is recommended. See the PCB Layout Guide-  
lines and Considerations section.  
1.0  
2.0  
3.0  
4.0  
10. Experience shows that the most frequent errors are using  
incorrect components, improper connections, and poor  
layout.  
5.0  
6.0  
7.0  
Performance Evaluation  
8.0  
This section shows the results of a random sample evalua-  
tion. Use these results as a reference guide for evaluating the  
RC5040 DC-DC converter for Pentium Pro motherboards.  
9.0  
10.0  
11.0  
12.0  
12.4  
Load Regulation 0.5A – 12.4A  
20  
APPLICATION NOTE  
AN42  
VID  
I
(A)  
V
(V)  
Low to High 0.5A-9.9A  
Current Step  
- 76.0mV  
+ 70mV  
Refer to  
load  
out  
Attachment  
A for Scope  
Picture  
1010  
0.5  
2.505  
2.504  
2.501  
2.496  
2.493  
2.493  
2.492  
2.492  
2.491  
2.490  
2.489  
2.488  
2.486  
2.485  
2.484  
0.84%  
1.0  
2.0  
High to Low 9.9A-0.5A  
Current Step  
Refer to  
Attachment  
B for Scope  
Picture  
3.0  
4.0  
5.0  
Low to High 0.5A-12.4A - 97.6mV  
Current Step  
Refer to  
6.0  
Attachment  
C for Scope  
Picture  
7.0  
8.0  
High to Low 12.4A-0.5A + 80.0mV  
Current Step  
Refer to  
9.0  
Attachment  
D for Scope  
Picture  
10.0  
11.0  
12.0  
13.0  
13.9  
Low to High 0.5A-13.9A - 99.2mV  
Current Step  
Refer to  
Attachment  
E for Scope  
Picture  
High to Low 13.9A-0.5A + 105.2mV Refer to  
Current Step  
Load Regulation 0.5 - 13.9A  
Attachment  
Note:  
F for Scope  
Picture  
Load regulation is expected to be typically around 0.8%. The  
load regulation performance for this device under evaluation is  
excellent.  
Note:  
Excellent transient voltage response. Transient voltage is rec-  
ommended to be less than 4% of the output voltage. The per-  
formance of the device under evaluation is significantly better  
than a typical VRM.  
Output Voltage LoadTransients Due to Load Current Step  
This test is performed using Intel P6.0/P6S/P6T Voltage  
Transient Tester.  
Input Ripple and Power on Input Rush Current  
Iload = 9.9A Input Ripple  
Refer to Attachment  
Voltage = 15mV G for Scope Picture  
Power on Input Rush Current is not measured on the mother-  
board because we did not want to cut the 5V trace and insert  
current probe in series with the supply. However, with the  
input filter design, the Input Rush Current will be well within  
specification.  
21  
AN42  
APPLICATION NOTE  
Component Case Temperature  
Case Temperature Case Temperature Case Temperature  
(°C)  
Iload= 9.9A  
(°C)  
Iload= 12.4A  
(°C)  
Iload =13.9A  
Device  
Description  
Q3A  
MOSFET  
K1388  
57  
58  
53  
66  
63  
64  
56  
70  
56.3  
66.6  
61.2  
87  
Q3B  
L1  
MOSFET  
K1388  
Inductor,  
Unknown  
Q2  
Schottky Diode  
2048CT  
IC  
Raytheon RC5040  
52  
38.2  
35  
54  
58  
39  
Cin  
Cout  
Input Capacitor 1000µF  
36.8  
34.8  
Output Capacitor  
38.2  
1500µF  
Note:  
Case temperatures are all within guidelines. Our guideline is that case temperatures for all components should be below 105°C  
@25°C Ambient.  
Comments:  
Excellent input ripple voltage. Input ripple voltage is recommended to be less than 5% of the output voltage.  
Evaluation Summary:  
The on-board DC-DC converter is fully functional. It has  
excellent load regulation, transient response, and input  
voltage ripple.  
Attachment A  
Attachment B  
22  
APPLICATION NOTE  
AN42  
Attachment E  
Attachment C  
Attachment F  
Attachment D  
Attachment G  
Summary  
RC5040/RC5042 Evaluation Board  
This application note covers for implementation of a DC-DC  
converter on a Pentium Pro motherboard using the RC5040  
and RC5042. The detailed discussion includes Pentium Pro  
processor power requirements, RC5040 and RC5042  
description, design considerationsn and component selec-  
tions, layout guidelines and considerations, guidelines for  
debugging, and performance evaluations.  
Raytheon Electronics provides an evaluation board for the  
purpose of verifying system level performance of the  
RC5040 and RC5042. The evaluation board serves as a guide  
as to what can be expected in performance with the supplied  
external components and PCB layout. Please call Raytheon  
Electronics Marketing Department at (415) 966-7819 for an  
evaluation board.  
23  
AN42  
APPLICATION NOTE  
Appendix A: Directory of Component Suppliers  
Dale Electronics, Inc.  
E. Hwy. 50, PO Box 180  
Yankton, SD 57078-0180  
PH: (605) 665-9301  
National Semiconductor  
2900 Semiconductor Drive  
Santa Clara, CA 95052-8090  
PH: (800) 272-9959  
Fuji Electric  
Nihon Inter Electronics Corp.  
Quantum Marketing Int’l, Inc.  
12900 Rolling Oaks Rd.  
Caliente, CA 93518  
Collmer Semiconductor Inc.  
14368 Proton Rd.  
Dallas, Texas 75244  
PH: (214)233-1589  
PH: (805) 867-2555  
General Instrument  
Panasonic Industrial Co.  
6550 Katella Avenue  
Cypress, CA 90630  
PH: (714) 373-7366  
Power Semiconductor Division  
10 Melville Park Road  
Melville, NY 11747  
PH: (516) 847-3000  
Pulse Engineering  
Hoskins Manufacturing Co.  
(Copel Resistor Wire)  
10776 Hall Road  
12220 World Trade Drive  
San Diego, CA 92128  
PH: (619) 674-8100  
Hamburg, MI 48139-0218  
PH: (313) 231-1900  
Sanyo Energy USA  
2001 Sanyo Avenue  
San Diego, CA 92173  
PH: (619) 661-6620  
Intel Corp.  
5200 NE Elam Young Pkwy.  
Hillsboro, OR. 97123  
PH: (800) 843-4481 Tech. Support  
for Power Validator  
Siliconix  
Temic Semiconductors  
2201 Laurelwood Road  
Santa Clara, CA 95056-1595  
PH: (800) 554-5565  
International Rectifier  
233 Kansas St.  
El Segundo, CA 90245  
PH: (310) 322-3331  
Sumida Electric USA  
5999 New Wilke Road Suite #110  
Rolling Meadows, IL 60008  
PH: (708) 956-0702  
IRC Inc.  
PO Box 1860  
Boone, NC 28607  
PH: (704) 264-8861  
Xicon Capacitors  
PO Box 170537  
Motorola Semiconductors  
PO Box 20912  
Arlington, Texas 76003  
PH:(800) 628-0544  
Phoenix, Arizona 85036  
PH:(602) 897-5056  
LIFE SUPPORT POLICY  
FAIRCHILD’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES  
OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR  
CORPORATION. As used herein:  
1. Life support devices or systems are devices or systems  
which, (a) are intended for surgical implant into the body,  
or (b) support or sustain life, and (c) whose failure to  
perform when properly used in accordance with  
instructions for use provided in the labeling, can be  
reasonable expected to result in a significant injury of the  
user.  
2. A critical component in any component of a life support  
device or system whose failure to perform can be  
reasonably expected to cause the failure of the life support  
device or system, or to affect its safety or effectiveness.  
Fairchild Semiconductor  
Corporation  
Fairchild Semiconductor  
Europe  
Fairchild Semiconductor  
Hong Kong Ltd.  
National Semiconductor  
Japan Ltd.  
Americas  
Fax: +49 (0) 1 80-530 85 86  
13th Floor, Straight Block,  
Ocean Center, 5 Canto Rd.  
Tsimshatsui, Kowloon  
Hong Kong  
Tel:81-3-5620-6175  
Fax:81-3-5620-6179  
Customer Response Center  
Tel:1-888-522-5372  
Deutsch Tel: +49 (0) 8 141-35-0  
English Tel: +44 (0) 1 793-85-68-56  
Italy  
Tel: +39 (0) 2 57 5631  
Tel:+852 2737-7200  
Fax:+852 2314-0061  
2/98 0.0m  
Stock#AN30000042  
1998 Fairchild Semiconductor Corporation  

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